Noise figure of active loop amplifiers – some thoughts

Review of noise

Let’s review of the concepts of noise figure, equivalent noise temperature and measurement.

Firstly let’s consider the nature of noise. The noise we are discussing is dominated by thermal noise, the noise due to random thermal agitation of charge carriers in conductors. Johnson noise (as it is known) has a uniform spectral power density, ie a uniform power/bandwidth. The maximum thermal noise power density available from a resistor at temperature T is given by \(NPD=k_B T\) where Boltzman’s constant kB=1.38064852e-23 (and of course the load must be matched to obtain that maximum noise power density). Temperature is absolute temperature, it is measured in Kelvins and 0°C≅273K.

Noise Figure

Noise Figure NF by definition is the reduction in S/N ratio (in dB) across a system component. So, we can write \(NF=10 log \frac{S_{in}}{N_{in}}- 10 log \frac{S_{out}}{N_{out}}\).

Equivalent noise temperature

One of the many methods of characterising the internal noise contribution of an amplifier is to treat it as noiseless and derive an equivalent temperature of a matched input resistor that delivers equivalent noise, this temperature is known as the equivalent noise temperature Te of the amplifier.

So for example, if we were to place a 50Ω resistor on the input of a nominally 50Ω input amplifier, and raised its temperature from 0K to the point T where the noise output power of the amplifier doubled, would could infer that the internal noise of the amplifier could be represented by an input resistor at temperature T. Fine in concept, but not very practical.

Y factor method

Applying a little maths, we do have a practical measurement method which is known as the Y factor method. It involves measuring the ratio of noise power output (Y) for two different source resistor temperatures, Tc and Th. We can say that \(NF=10 log \frac{(\frac{T_h}{290}-1)-Y(\frac{T_c}{290}-1)}{Y-1}\).

AN 57-1 contains a detailed mathematical explanation / proof of the Y factor method.

We can buy a noise source off the shelf, they come in a range of hot and cold temperatures. For example, one with specified Excess Noise Ratio (a common method of specifying them) has Th=9461K and Tc=290K. If we measured a DUT and observed that Y=3 (4.77dB) we could calculate that NF=12dB. Continue reading Noise figure of active loop amplifiers – some thoughts

Small untuned loop for receiving – a design walk through #3

Small untuned loop for receiving – a design walk through #1 arrived at a design concept comprising an untuned small loop loaded with a broadband amp with input Z being a constant resistive value and with frequency independent gain and noise figure.

Small untuned loop for receiving – a design walk through #2 developed a simple spreadsheet model of the loop in free space loaded by the amplifier andperformed some basic SND calculations arriving at a good candidate to take to the next stage, NEC modelling.

The simple models previously used relied upon a simple formula for predicting radiation resistance Rr in free space, and did not capture the effects of proximity of real ground. The NEC model will not be subject to those limitations, and so the model can be run from 0.5-30MHz.

The chosen geometry was:

  • loop perimeter: 3.3m;
  • conductor diameter: 20mm;
  • transformer ratio to 50Ω amplifier: 0.7; and
  • height of the loop centre: 2m;
  • ground: average (σ=0.005 εr=13).

NEC-5.0 model results

The effect of interaction with nearby real ground is to modify the free space radiation pattern. The pattern at low frequencies has maximum gain at the zenith, and above about 15MHz the pattern spreads and maximum gain is at progressively lower elevation. For the purposes of a simple comparison, the AntennaFactor was calculated for external plan wave excitation at 45° elevation in the plane of the loop.

Above is a plot of loop Gain and AntennaFactor at 45° elevation along the loop axis. The frequency range is 0.5-30MHz as the NEC model is not limited by the simple Rr formula. Additionally there is some ‘ground gain’ of around 5dB due to lossy reflection of waves from the ground interface. Continue reading Small untuned loop for receiving – a design walk through #3

Small untuned loop for receiving – a design walk through #2

Small untuned loop for receiving – a design walk through #1 arrived at a design concept comprising an untuned small loop loaded with a broadband amp with input Z being a constant resistive value and with frequency independent gain and noise figure.

Loop amplifier

There have been many credible designs of loop amplifiers of gain in the region of 25+dB and NoiseFigure NF around 2dB. So lets work with that as a practical type of amplifier, though we will not commit to input Z just yet.

I might note that a certain active loop manufacturer claims NF in the small tenths of a dB, but it appears they needed to invent their own method of measurement… when questions the credibility of their claims.

Let’s calculate the NF of a cascade of the NF=8dB receiver, coax with loss of 2dB and a loop amplifier with NF=2dB and Gain=25dB.

The NF looking into the loop amplifier is 2.08dB. Continue reading Small untuned loop for receiving – a design walk through #2

Small untuned loop for receiving – a design walk through #1

This series of articles develops a simple design for a small receive only broadband loop for the frequency range 0.5-10MHz, and to deliver fairly good practical sensitivity.

Fairly good practical sensitivity is to mean that the recovered S/N ratio is not much worse than the off-air S/N ratio. Let’s quantify not much worse as the Signal to Noise Degradation (SND) statistic calculated as \(SND=10 log\frac{N_{int}+N_{ext}}{N_{ext}}\), and lets set a limit that \(SND<3 dB\).

Since Next is part of the criteria, let’s explore it.

External noise

ITU-R P.372 gives us guidance on the expected median noise levels in a range of precincts. Since most hams operate in residential areas, you might at first think the Residential precint is the most appropriate, but ambient noise more like the Rural precinct is commonly observed in residential areas, so let’s choose Rural as a slightly ambitious target.

Above is Fig 39 from ITU-R P.372-14 showing the ambient noise figure for the range of precincts. Readers will not that that are all lines sloping downwards with increasing frequency, so the external noise floor is greater at lower frequencies in this range. Continue reading Small untuned loop for receiving – a design walk through #1

Review of iFIX RT300M v2 LED tester

In other posts on LED luminairs, I identified the need for a test device for LED strings of up to 200+V at currents up to 280mA.

There are quite a number of competitive devices in the market, the article is a review of the iFIX RT300M v2 (which is also sold under other brand names, they may or may not be sourced from the same factory… the Chinese are copyists).

 

I purchased one of these devices, and it was faulty on delivery. The output voltage never rises above 0.3V, determined to be a hardware fault. With eBay’s intervention, a full refund was obtained without returning the faulty unit which turned out to be a small blessing later.

The v2 RT300M has a button on the top edge of the device. Continue reading Review of iFIX RT300M v2 LED tester

Finding the electrical length of the branches of an N type T – #4

Finding the electrical length of the branches of an N type T – #1 posed a problem, this article looks at one solution.

For convenience, here is the problem.

An interesting problem arises in some applications in trying to measure the electrical length of each branch of a N type T piece.

Let’s make some assumptions that the device is of quality, that the connection from each connector to the internal junction is a uniform almost lossless transmission line of Zo=50Ω. Don’t assume that the left and right branches above are of the same length (though they often are) and we should not assume that the nearest branch is of the same length as the others (and they are often not).

Before we start, we will calibrate the VNA entering the offsets appropriate to the OPEN and SHORT cal parts. In the case of my nanoVNA, measurement above 900MHz is very noisy, so the scan will be 100-890MHz (to avoid a glitch at 900MHz due to harmonic mode switching).

So, the problem is all the uncertain things that connect to the internal T junction. Lets connect a calibration quality 50Ω termination to the left hand port. We now know that the path from the male port to the remaining female port comprises lengths l2 and l1 of low loss 50Ω transmission line with a 50Ω resistor shunting at the junction of l1 and l2. Continue reading Finding the electrical length of the branches of an N type T – #4

Finding the electrical length of the branches of an N type T – #3

Finding the electrical length of the branches of an N type T – #1 posed a problem, this article looks at one solution.

For convenience, here is the problem.

An interesting problem arises in some applications in trying to measure the electrical length of each branch of a N type T piece.

Let’s make some assumptions that the device is of quality, that the connection from each connector to the internal junction is a uniform almost lossless transmission line of Zo=50Ω. Don’t assume that the left and right branches above are of the same length (though they often are) and we should not assume that the nearest branch is of the same length as the others (and they are often not).

Before we start, we will calibrate the VNA entering the offsets appropriate to the OPEN and SHORT cal parts.

So, the problem is all the uncertain things that connect to the internal T junction. Lets connect a calibration quality 50Ω termination to the left hand port. We now know that the path from the male port to the remaining female port comprises lengths l2 and l1 of low loss 50Ω transmission line with a 50Ω resistor shunting at the junction of l1 and l2. Continue reading Finding the electrical length of the branches of an N type T – #3

Finding the electrical length of the branches of an N type T – #2

Finding the electrical length of the branches of an N type T – #1 posed a problem, this article looks at one simple solution.

For convenience, here is the problem.

An interesting problem arises in some applications in trying to measure the electrical length of each branch of a N type T piece.

Let’s make some assumptions that the device is of quality, that the connection from each connector to the internal junction is a uniform almost lossless transmission line of Zo=50Ω. Don’t assume that the left and right branches above are of the same length (though they often are) and we should not assume that the nearest branch is of the same length as the others (and they are often not).

So, the problem is all the uncertain things that connect to the internal T junction. Lets connect a calibration quality 50Ω termination to the left hand port. We now know that the path from the male port to the remaining female port comprises lengths l2 and l1 of low loss 50Ω transmission line with a 50Ω resistor shunting at the junction of l1 and l2. Continue reading Finding the electrical length of the branches of an N type T – #2

Finding the electrical length of the branches of an N type T – #1

An interesting problem arises in some applications in trying to measure the electrical length of each branch of a N type T piece.

Let’s make some assumptions that the device is of quality, that the connection from each connector to the internal junction is a uniform almost lossless transmission line of Zo=50Ω. Don’t that the left and right branches above are of the same length (though they often are) and we should not assume that the nearest branch is of the same length as the others (and they are often not).

Something to keep in mind is that the reference plane for the female connectors is about 9mm inside the T, and you can see the reference plane on the male connector, the nearest end of the shield connection.

With a ruler, the physical length of the left and right female branches looks to be about 13mm, and around 28mm for the male branch… but electrical length will be longer due to an unknown (as yet) deployment of dielectric of unknown type inside the T.

So, put your thinking caps on.

A solution to follow…

Performance of a small transmitting loop with varying height – NEC-5.0

Around 2015 I constructed a series of models exploring the effect of ground proximity on a small transmitting loop (STL).

At frequency 7.2MHz, the loop was octagonal with area of 1m^2 equivalent radius a=0.443m, ka=0.067rad, 3.15mm radius copper conductor, lossless tuning capacitor, and centre height above ground (σ=0.007  εr=17 ) was varied from 1.5 to 10m (0.036-0.240λ).

The model series was run in NEC-2, NEC-4.1, NEC-4.2 and NEC-5.0, and the results varied. NEC-4.1 showed serious problems, eg negative input resistance at some heights. The problem was discussed the Burke, and he explained that there was a known problem in NEC-4.1 for small loops near ground, and sent me an upgrade to NEC-4.2 to try with the GN 3 ground model, but that the better solution was in NEC-5 if it was ever released.

NEC-4.2 solved the negative resistance problem, but some issues remained.

With the recent release of NEC-5.0, opportunity arises to compare all four approaches.

(Burke 2019) p45 discusses loop antennas over ground and NEC-5.0.

The plot above of radiation efficiency gives an overall comparison of the different model techniques. (Burke 2019) states Since the mixed-potential solution ensures that the approximated integral of scalar potential around the loop is zero, whether the potential is accurate or not, it might be expected to do better than NEC-4. Continue reading Performance of a small transmitting loop with varying height – NEC-5.0