## nanoVNA-H – sweep of a coax line section with OC termination

This article discusses the use of the (modified) nanoVNA-H raw accuracy and the implications for calibrated measurements.

## Introduction

VNAs achieve much of their accuracy by applying a set of error corrections to a measurement data set.

The error corrections are obtained by making ‘raw’ measurements of a set of known parts, most commonly a short circuit, open circuit and load resistor (the OSL parts). The correction data may assume each of these parts is ideal, or it may provide for inclusion of a more sophisticated model of their imperfection. This process is known as calibration of the instrument and test fixture. nanovna-Q appears to have some fixed departure compensation to suit the SMA cal parts, less suited to other test fixtures.

So, when you make a measurement at some frequency, the correction data for THAT frequency is retrieved and used to correct the measurement.

What if there is not correction data for THAT frequency? There are two approaches:

• a calibration run is required for exactly the same frequency range and steps (linear, logarithmic, size) as the intended measurement; and
• existing calibration data is interpolated to the frequency of interest.

The interpolation method is convenient, but adds uncertainty (error) to the measurement. Some commercial VNAs will NOT interpolate.

The nanoVNA will interpolate, and with interpolation comes increased uncertainty.

An uncorrected sweep of a reasonably known DUT is revealing of the instrument inherent error.

The DUT is a 12m length of LMR400.

## Expected behavior

Let’s first estimate how it should behave.

The VNA contains a directional coupler nominally designed / calibrated for Zo=50+j0Ω, and in use, VNAs are invariably used to display measurements in terms of some purely real impedance, commonly 50Ω.

Though the DUT characteristic impedance (Zo) is nominally 50Ω, it is not EXACTLY 50+j0Ω and so there are departures in the displayed values wrt 50Ω from what might happen in terms of the actual Zo.

We can calculate the magnitude of Gamma for our 12m OC section of LMR400 over a range of frequencies. |Gamma| vs frequency is a smooth curve as a result of line attenuation increasing with frequency. As a result in the small departure in Zo, |Gamma| wrt 50Ω has a superimposed small decaying oscillation. Continue reading nanoVNA-H – sweep of a coax line section with OC termination

## nanoVNA-H – T-Check test

Rhode & Schwarz describe a test for accuracy of a VNA at T-Check Accuracy Test for Vector Network Analyzers utilizing a Tee-junction.

A nanoVNA-H PCB v3.3 (modified to fix decoupling problem on mixers) was calibrated from 0.1-900MHz using the supplied parts.

A T piece with extra 50Ω termination was inserted between the supplied original cables and s11 and s21 captured. The assembly was turned around and measured again to capture s22 and s12 (though recorded as s11 and s21). The two files were merged to obtain a full two port bothways .s2p file.

The T-Check value was calculated and is plotted here in VNWA. Above, the T-Check results are not stunning at all, the ideal result is 1.0 at all frequencies. Rhode and Schwarz recommend that more than 15% error is unacceptable… of course that is in a commercial grade VNA.

## nanoVNA-H – measure ferrite transformer

This article demonstrates the use of the (modified) nanoVNA-H to measure Loss (Transmission Loss) and Insertion Loss of a small ferrite 64:1 RF transformer, and the Insertion VSWR and Return Loss. The transformer was designed for a receive application at 9MHz.

Firstly let’s define the meaning of the terms: Continue reading nanoVNA-H – measure ferrite transformer

## nanoVNA-H – measure ferrite core permeability

This article demonstrates the use of the (modified) nanoVNA-H to capture data from which the complex relative permeability of an unknown ferrite core is calculated and plotted Above, a single turn of wire through the sleeve allows measurement by the nanovna. The nanoVNA fundamentally captures s11 parameters which we need to convert to relative permeability. Continue reading nanoVNA-H – measure ferrite core permeability

## nanoVNA-H – measure equivalent core loss resistance

A very common design of a n:1 transformer for EFHW antennas uses a 2t primary on and FT240-43 (or even smaller) ferrite core.

In a process of designing a transformer, we often start with an approximate low frequency equivalent circuit. “Low frequency” is a relative term, it means at frequencies where each winding current phase is uniform, and the effects of distributed capacitance are insignificant. Above is a commonly used low frequency equivalent of a transformer. Z1 and Z2 represent leakage impedances (ie the effect of magnetic flux leakage) and winding conductor loss. Zm is the magnetising impedance and represents the self inductance of the primary winding and core losses (hysteresis and eddy current losses). Continue reading nanoVNA-H – measure equivalent core loss resistance

## Designing with binocular ferrite cores – published Al values

Designing with ferrite binocular cores can be frustrating as there are different formats in which data is provided, and data for different mixes on the same dimensioned cores appear inconsistent.

There are several source of the Al parameter for some common cores, often from resellers rather than manufacturers. Continue reading Designing with binocular ferrite cores – published Al values

## Designing with binocular ferrite cores – Σ(A/l)

Designing with ferrite binocular cores can be frustrating as there are different formats in which data is provided, and data for different mixes on the same dimensioned cores appear inconsistent.

## #61 mix

This article documents calculated geometry Σ(A/l) derived for a number of Fair-rite cores from their specified Al (at µi). Continue reading Designing with binocular ferrite cores – Σ(A/l)

## RF transformer design with ferrite cores – saturation calcs

Ferrite cored inductors and transformers saturate at relatively low magnetising force.

## #61 material example

Lets work through an example of a FT50-61 core with 10t primary at 3.5MHz.

Magnetic saturation is one limit on power handling capacity of such a transformer, and likely the most significant one for very low loss cores (#61 material losses are very low at 3.5MHz).

Let’s calculate the expected magnetising impedance @ 3.5MHz.

## A review of transformer design

In a process of designing a transformer, we often start with an approximate low frequency equivalent circuit. “Low frequency” is a relative term, it means at frequencies where each winding current phase is uniform, and the effects of distributed capacitance are insignificant. Above is a commonly used low frequency equivalent of a transformer. Z1 and Z2 represent leakage impedances (ie the effect of magnetic flux leakage) and winding conductor loss. Zm is the magnetising impedance and represents the self inductance of the primary winding and core losses (hysteresis and eddy current losses). Continue reading RF transformer design with ferrite cores – initial steps