This article canvasses a possible improvement of the s21 shunt-through measurement of impedance to compensate for errors in VNA port impedances that are not corrected in simpler calibration / correction schemes.
The diagram above is from (Agilent 2009) and illustrates the configuration of a shunt-through impedance measurement. Continue reading Improving ‘s21 shunt-through’ measurement of low impedances
With the popularity of the nanoVNA, the matter of optimisation of antenna systems comes up and the hoary chestnuts of ham radio are trotted out yet again.
Having skimmed a presentation published on the net, an interesting example is presented of an 80m half wave centre dipole with feed line and various plots from the nanoVNA used to illustrate the author’s take on things.
The author is obsessed with resonance and obsessed with phase, guiding the audience to phase as ‘the’ optimisation target. Phase of what you might ask… all the plots the author used to illustrate his point are phase of s11.
I have constructed an NEC-4.2 model of a somewhat similar antenna to illustrate sound concepts. Since NEC-4.2 does not model lossy transmission lines (TL elements), we will import the feed point data into Simsmith to include transmission line loss in the model.
Above is the Simsmith model. Continue reading Antenna system resonance and the nanoVNA
Improving ‘s21 series-through’ measurement of high impedances canvassed a possible improvement of the s21 series-through measurement of impedance to compensate for errors in VNA port impedances that are not corrected in simpler calibration / correction schemes.
This article provides more detail on the practical test case.
A small ferrite cored test inductor was measured with a ‘bare’ nanoVNA SOLT calibrated, firstly using s11 reflection.
Above is the R,X,|Z| plot from the s11 reflection measurement of the unknown Zu. Continue reading Improving ‘s21 series-through’ measurement of high impedances – more detail
This article canvasses a possible improvement of the s21 series-through measurement of impedance to compensate for errors in VNA port impedances that are not corrected in simpler calibration / correction schemes.
The diagram above is from (Agilent 2009) and illustrates the configuration of a series-through impedance measurement. Continue reading Improving ‘s21 series-through’ measurement of high impedances
I often need to calculate loss from marker values on a VNA screen, or extracted from a saved .s2p file.
Firstly, loss means PowerIn/PowerOut, and can be expressed in dB as 10log(PowerIn/PowerOut). For a passive network, loss is always greater than unity or +ve in dB.
\(loss=\frac{PowerIn}{PowerOut}\\\)Some might also refer to this as Transmission Loss to avoid doubt, but it is the fundamental meaning of loss which might be further qualified.
So, lets find the two quantities in the right hand side using ‘powerwaves’ as used in S parameter measurement.
s11 and s21 are complex quantities, both relative to port 1 forward power, so we can use them to calculate relative PowerIn and relative PowerOut, and from that PowerIn/PowerOut.
PowerIn is port 1 forward power less the reflected power at port 1, \(PowerIn=P_{fwd} \cdot (1-|s11|^2)\).
PowerOut is port 2 forward power times less the reflected power at the load (which we take to be zero as under this test it is a good 50Ω termination), \(PowerOut=P_{fwd} \cdot |s21|^2 \).
So, we can calculate \(loss=\frac{PowerIn}{PowerOut}=\frac{\frac{PowerIn}{P_{fwd}}}{ \frac{PowerOut}{P_{fwd}}}=\frac{1-|s11|^2}{|s21|^2}\)
Noelec makes a small transformer, the Balun One Nine, pictured above and they offer a set of |s11| and |s12| curves in a back to back test. (Note: back to back tests are not a very reliable test.) Continue reading Calculate Loss from s11 and s21 – convenient online calculator
There are many ways to get a good estimate of the characteristic impedance Zo of a transmission line.
One method is to measure the input impedances of a section of line with both a short circuit and open circuit termination. From Zsc and Zoc we can calculate the Zo, and the complex propagation constant \(\gamma=\alpha + \jmath \beta\).
Calculation of Zo is quite straightforward.
The solution for γ involves the log of a complex number \(r \angle \theta\) which is one of the many possible values \(ln(r) + j \left(\theta + 2 \pi k \right)\) for +ve integer k. Conveniently, the real part α is simply \(ln(r) \). The real part of γ is the attenuation in Np/m which can be scaled to dB/m, and the imaginary part is the phase velocity in c/m. The challenge is finding k.
So, let’s measure a sample of 14×0.14, 0.22mm^2, 0.5mm dia PVC insulated small speaker twin.
Above is the nanoVNA setup for measurement. Continue reading Measure transmission line Zo – nanoVNA – PVC speaker twin
There are many ways to get a good estimate of the characteristic impedance Zo of a transmission line.
One method is to measure the input impedances of a section of line with both a short circuit and open circuit termination. From Zsc and Zoc we can calculate the Zo, and the complex propagation constant \(\gamma=\alpha + \jmath \beta\).
Calculation of Zo is quite straightforward.
The solution for γ involves the log of a complex number \(r \angle \theta\) which is one of the many possible values \(ln(r) + j \left(\theta + 2 \pi k \right)\) for +ve integer k. Conveniently, the real part α is simply \(ln(r) \). The real part of γ is the attenuation in Np/m which can be scaled to dB/m, and the imaginary part is the phase velocity in c/m. The challenge is finding k.
Let’s take an example from recent measurements of 35m of CCS RG6 coax, and extract the s11 values recorded in saved .s1p files @ 1.87MHz. The saved data in MA format, magnitude and angle (in degrees).
Calculate Zo and gamma is flexible and can accept the MA format data directly.
Above, the results. Zo is 74.73-j1.156Ω, and matched line loss MLL is 0.03281dB/m. This MLL is quite a deal higher than you might find in many line loss calculators, they often fail on CCS cables. Continue reading Measure transmission line Zo – nanoVNA – CCS RG6
My nanoVNA-H v3.3 is just over six months old. It has succumbed to most of the common hardware faults and some of the less common ones, but until now the touch screen has worked ok.
That has changed, the bezel bears on the touch screen and causes input when the case is held even very lightly by the edges… sufficiently so that it was unusable.
A perhaps temporary resolution was to place four 1mm thick M2 5mm OD nylon washers on the front part of the case before carefully placing the PCB assembly on top, then the case back and screws. This is a bit tedious if one of the washers moves, next time it is apart to fix hardware issues, I might put a dot of CA adhesive under each washer.
The case design is inadequate, it needed some features incorporated into the molding of the front to space the edges of the bezel off the PCB so that it maintained the requisite spacing even when held, even after the molding changes over time (perhaps normalisation of stresses from the molding process).
Fair-rite allocates some of its closed loop ferrite products to two different categories:
Sometimes the same dimensioned cores are available in both categories with different part numbers and possibly different prices, implying some real difference in behavior, eg 5943003801 and 2643803802 are both FT240-43 sized cores.
Material datasheets often contain a note like this from the #43 datasheet:
Characteristic curves are measured on standard Toroids (18/10/6 mm) at 25°C and 10 kHz unless otherwise indicated. Impedance characteristics are measured on standard shield beads (3.5/1.3/6.0 mm) unless otherwise indicated.
I sought to clarify my interpretation of this clause by asking Fair-rite …whether the published material permeability curves / tables apply to suppression product. Can I use the published permeability curves / tables to predict inductor impedance reliably for suppression products?
Fair-rite’s Michael Arasim advised… Continue reading Using complex permeability to design with Fair-rite suppression products
Some of my articles have contained thermal pictures of ferrite cored inductors and transformers.
I have been asked several times recently about the assumed emissivity and the accuracy questioned, I assume this has been discussed online somewhere.
When first measuring ferrites with non-contact thermometers, I performed some experiments to discover whether the default emissivity ε=0.95 applied. It would be convenient if it did, and permit use of some instruments that do not allow adjustment of ε.
In the past, I have compared the reading of non-contact thermometers with several K thermocouple meters and a Thermomelt indicator, and observed insignificant difference (ie less than the variance of repeated measurements).
The following experiment is a thermal pic of a FT240-43 core on the black plastic case of the instrument. The setup has had hours to stabilise thermally.
Above is a combined thermal image and faint visual image. This instrument has only one readout point, and by moving it around, only 0.1° variation was observed between the background and the core. Continue reading Black body emissivity of ferrite core material