## nanoVNA – evaluation of a voltage balun – W2AU 1:1

In this article, I will outline an evaluation of a ‘classic’ voltage balun, the 1:1 W2AU voltage balun, specified for 1.8-30MHz.

These were very popular at one time, but good voltage baluns achieve good current balance ONLY on very symmetric loads and so are not well suited to most wire antennas.

Above is W2AU’s illustration of the internals.

Mine barely saw service before it became obvious that it had an intermittent connection to the inner pin of the coax connector. That turned out to be a poor soldered joint, a problem that is apparently quite common and perhaps the result of not properly removing the wire enamel before soldering.

Having cut the enclosure to get at the innards and fix it (they were not intended to be repaired), I rebuilt it in a similar enclosure made from plumbing PVC pipe and caps, and took the opportunity to fit some different output terminals and an N type coax connector.

Above is the rebuilt balun which since that day has been reserved for test kit for evaluating the performance of a voltage balun in some scenario or another. Continue reading nanoVNA – evaluation of a voltage balun – W2AU 1:1

## nanoVNA – RG6/U with CCS centre conductor MLL measurement

In my recent article RG6/U with CCS centre conductor – shielded twin study I made the point that it is naive to rely upon most line loss calculators for estimating the loss of this cable type partly because of their inability to model the loss at low HF and partly because of the confidence one might have in commonly available product. In that article I relied upon measured data for a test line section.

I have been asked if the nanoVNA could be bought to bear on the problem of measuring actual matched line loss (MLL). This article describes one method.

The nanoVNA has been OSL calibrated from 1-299MHz, and a 35m section of good RG6 quad shield CCS cable connected to Port 1 (Ch0 in nanoVNA speak).

A sweep was made from 1-30MHz with the far end open and shorted and the sweeps saved as .s1p files.

Above is a screenshot of one of the sweeps. Continue reading nanoVNA – RG6/U with CCS centre conductor MLL measurement

## RG6/U with CCS centre conductor – shielded twin study

Some online experts advise the use of synthesised shielded twin instead of ordinary two wire line for HF antennas claiming it is vastly superior.

Now it could be vastly superior for several reasons in all, but let’s focus on just one important parameter, loss under mismatch conditions.

The scenario then is the very popular 132′ multi band dipole:

• the famous 40m (132′) centre fed dipole;
• 20m of feed line being parallel RG6/U CCS quad shield with shields bonded at both ends;
• 7MHz where we will assume dipole feed point impedance is ~4000+j0Ω.

We will consider the system balanced and only deal with differential currents.

Now rather than depend on loss calculators, most of which don’t reconcile with measurement of CCS RG6/U, I will used measured loss. RG6/U with CCS centre conductor at HF gives a chart of measured loss of a sample of commercial grade CCS quad shield coax.

Above is a comparison of matched line loss (MLL) based on measurement of a length of RG6/U Quad Shield CCS cable and prediction from Simsmith of Belden 8215 (also CCS). The ripple is due to measurement system error, measurements were made quite some years ago with a AIMuhf. Continue reading RG6/U with CCS centre conductor – shielded twin study

## A dipole centre insulator from HDPE cutting board

I made a small dipole centre insulator from 10mm HDPE cutting board on the CNC router. HDPE is moderately UV resistant so should survive for some years outdoors.

The insulator is 100mm across its widest points. No provision is made for supporting coax, it is for use with home made open wire line which will fall from the dipole leg ends. If you want to use it with coax or ribbon feedline, then incorporate a tab to secure those lines.

## A model of current distribution in copper clad steel conductors at RF – capturing conductor curvature

A model of current distribution in copper clad steel conductors at RF laid out a model for current distribution, though ignoring curvature of the conductor in calculating current density vs depth.

A model for current distribution in a conductor is that for a homogenous conducting half space with surface current parallel to the interface. Current density at depth d is given by the expression $$J_r=J_R\frac{J_0(kr)}{J_0(kR)}$$ where δ is the skin depth $$δ=(ω \cdot µ \cdot σ)^{0.5}$$ and $$k=\frac{1-\jmath}{\delta}$$, σ is the conductivity). This takes into account curvature of the conductor surface, albeit with slower compute time.

Let’s compare the two algorithms on a test case at 1.8MHz being copper cladding of 67µm copper over a steel core for an overall diameter of 1.024mm (#18).

Above is a stacked image, the simpler algorithm is the feint plot.

There is a quite small difference in this case. When the expected loss of 400Ω line using the conductor is calculated, the result with the simpler algorithm is 1.3% less than the later one using the Bessel distribution.

## A Smith chart view of EFHW transformer compensation

I have written several articles on design of high ratio ferrite cored transformers for EFHW antennas.

Having selected a candidate core, the main questions need to be answered:

• how many turns are sufficient for acceptable InsertionVSWR at low frequencies and core loss; and
• what value of shunt capacitance best compensates the effect of leakage inductance at high frequencies?

Lets look at a simplified equivalent circuit of such a transformer, and all components are referred to the 50Ω input side of the transformer.

Above is a simplified model that will illustrate the issues. For simplicity, the model is somewhat idealised in that the components are lossless. Continue reading A Smith chart view of EFHW transformer compensation

## Stacking of Yagis and antenna effective aperture

The matter of stacking Yagis for improved gain is it seems a bit of a black art (and it should not be).

A common piece of advice is to visualise the capture area of the individual Yagi, and to stack them so that their capture areas just touch… with the intimation that if they overlap, then significant gain is lost.

Above is a diagram from F4AZF illustrating the concept. Similar diagrams exist on plenty of web sites, so it may not be original to F4AZF. Continue reading Stacking of Yagis and antenna effective aperture

## Desk review of the AAA-1C as an active dipole antenna

The AAA-1C is an amplifier for small receiving antennas by LZ1AQ. The amplifier is designed for use with one or two small loops or a short dipole (possibly comprising two small loops).

The datasheet contains some specifications that should allow calculation of S/N degradation (SND) in a given ambient noise context (such as ITU-R P.372). Of particular interest to me is the frequency range 2-30MHz, but mainly 2-15MHz.

The specifications would appear to be based on models of the active antenna in free space, or measurements of the device using a dummy antenna. So, the challenge is to derive some equivalent noise estimates that can be compared to P.372 ambient noise, and with adjustment for the likely effects of real ground.

Key specifications:

• plot of measured output noise of the amplifier, and receiver noise in 1kHz ENB;
• Antenna Factor (AF) from a simulation.

Above is the published noise measurements at the receiver input terminals. The graph was digitised and then a cubic spline interpolation used to populate a table. Continue reading Desk review of the AAA-1C as an active dipole antenna

## Small untuned loop for receiving – a design walk through #4

Small untuned loop for receiving – a design walk through #1 arrived at a design concept comprising an untuned small loop loaded with a broadband amp with input Z being a constant resistive value and with frequency independent gain and noise figure.

In that instance, the design approach was to find a loop geometry that when combined with a practical amplifier of given (frequency independent) NoiseFigure (NF), would achieve a given worst case S/N degradation (SND). Whilst several options for amplifier Rin were considered in the simple analytical model, the NEC mode of the antenna in presence of real ground steered the design to Rin=100Ω.

A question that commonly arises is that of Rin, there being two predominant schools of thought:

• Rin should be very low, of the order of 2Ω; and
• Rin should be the ‘standard’ 50Ω.

Each is limiting… often the case of simplistic Rules of Thumb (RoT).

Let’s plot loop gain and antenna factor for two scenarios, Rin=2Ω and Rin=100Ω (as used in the final design) from the simple model of the loop used at Small untuned loop for receiving – a design walk through #2.

Above, loop gain is dominated by the impedance mismatch between the source with Zs=Rr+Xl and the load being Rin. We can see that the case of Rin=100Ω achieves higher gain at the higher frequencies by way of less mismatch loss than the Rin=2Ω case. Continue reading Small untuned loop for receiving – a design walk through #4

## Small untuned loop for receiving – a design walk through #3

Small untuned loop for receiving – a design walk through #1 arrived at a design concept comprising an untuned small loop loaded with a broadband amp with input Z being a constant resistive value and with frequency independent gain and noise figure.

Small untuned loop for receiving – a design walk through #2 developed a simple spreadsheet model of the loop in free space loaded by the amplifier andperformed some basic SND calculations arriving at a good candidate to take to the next stage, NEC modelling.

The simple models previously used relied upon a simple formula for predicting radiation resistance Rr in free space, and did not capture the effects of proximity of real ground. The NEC model will not be subject to those limitations, and so the model can be run from 0.5-30MHz.

The chosen geometry was:

• loop perimeter: 3.3m;
• conductor diameter: 20mm;
• transformer ratio to 50Ω amplifier: 0.7; and
• height of the loop centre: 2m;
• ground: average (σ=0.005 εr=13).

## NEC-5.0 model results

The effect of interaction with nearby real ground is to modify the free space radiation pattern. The pattern at low frequencies has maximum gain at the zenith, and above about 15MHz the pattern spreads and maximum gain is at progressively lower elevation. For the purposes of a simple comparison, the AntennaFactor was calculated for external plan wave excitation at 45° elevation in the plane of the loop.

Above is a plot of loop Gain and AntennaFactor at 45° elevation along the loop axis. The frequency range is 0.5-30MHz as the NEC model is not limited by the simple Rr formula. Additionally there is some ‘ground gain’ of around 5dB due to lossy reflection of waves from the ground interface. Continue reading Small untuned loop for receiving – a design walk through #3