High end VSWR compensation in a ferrite cored RF transformer

The article Estimating the Insertion VSWR in a ferrite cored RF transformer discussed the importance of sufficient magnetising impedance to InsertionVSWR at low frequencies.

Above is a low frequency equivalent circuit of a transformer. Although most accurate at low frequencies, it is still useful for RF transformers but realise that it does not include the effects of distributed capacitance which have greater effect with increasing frequency.

The elements r1,x1 and r2,x2 model winding resistance and flux leakage as an equivalent impedance. Whilst for low loss cores at power frequencies, flux leakage is thought of as an equivalent inductance, purely reactive and proportional to frequency, the case of lossy ferrite cores at RF is more complicated. Winding resistance with well developed skin effect increases proportional to the square of frequency, but with lossy ferrite cores will often be dwarfed by the loss element of leakage impedance. Continue reading High end VSWR compensation in a ferrite cored RF transformer

Estimating the Insertion VSWR in a ferrite cored RF transformer

The article Estimating the magnetising or core loss in a ferrite cored RF transformer discussed a first cut approach to determining the minimum magnetising impedance from a core loss viewpoint.

This article considers the effect of magnetising impedance on VSWR.

For medium to high µ cored RF transformers, flux leakage should be fairly low and the transformer can be considered to be an ideal transformer of nominal turns ratio shunted at the input by the magnetising impedance observed at that input winding.

A good indication of the nominal impedance transformation of the combination is to find the VSWR of the magnetising impedance in shunt with the nominal load (eg 50+j0Ω in many cases), and to express this as InsertionVSWR when the transformer is loaded with a resistance equal to n^2*that nominal load (eg 50+j0Ω in many cases). This model is better for low values of n than higher, but it can still provide useful indication for n as high as 8 if flux leakage is low.

Magnetising impedance can be estimated using one of the following calculators, but keep in mind that there are quite wide tolerances on ferrite cores.

Magnetising impedance can be measured (eg with an analyser), but it should be measured with only the measured winding on the core. Did I mention the wide tolerance of ferrites?

Example – FT240-43 3t @ 3.6MHz

You might ask the question is 3t sufficient for the primary of an EFHW transformer that delivers a 50+j0Ω load to a transmitter. Continue reading Estimating the Insertion VSWR in a ferrite cored RF transformer

Estimating the magnetising or core loss in a ferrite cored RF transformer

The article End fed matching – design review and many later ones set out a method of estimating the magnetising or core loss in a ferrite cored RF transformer (such as often used with EFHW antennas).

There are two elements that are critical to efficient near ideal impedance transformation over a wide frequency range, low flux leakage and sufficiently high magnetising impedance. While low magnetising loss is essential for efficiency, it does not guarantee sufficiently high magnetising impedance for near ideal impedance transformation.

Magnetising impedance can be estimated using one of the following calculators, but keep in mind that there are quite wide tolerances on ferrite cores.

Magnetising impedance can be measured (eg with an analyser), but it should be measured with only the measured winding on the core. Did I mention the wide tolerance of ferrites?

Example – FT240-43 3t @ 3.6MHz

You might ask the question is 3t sufficient for the primary of an EFHW transformer that delivers a 50+j0Ω load to a transmitter. Continue reading Estimating the magnetising or core loss in a ferrite cored RF transformer

Derating rectifier diode current

The ratings for rectifier diode currents are often expressed in terms of a steady (ie DC) current, yet when used as a power rectifier with capacitor input filters, the current is very different.

Above is capture of the rectifier input current for a lab power supply set to 1A load current, the scale factor for the current probe is 1V/A. Continue reading Derating rectifier diode current

Using SPICE on antenna baluns

Guanella’s 1:1 balun and his explanation gave a LTSPICE model of Guanella’s 1:1 balun.

The LTSPICE model was of a ‘test bench’ implementation of the balun which comprised an air cored solenoid of two wire transmission line, with a slightly asymmetric lumped load.

This article discusses limitations of SPICE in modelling practical baluns.

Guanella’s 1:1 balun and his explanation – Zcm gave the characteristics of a example ferrite cored balun.

Above is Zcm of a 11t balun wound on a FT240-43 toroid. The ferrite core acts on the common mode choke element and has negligible effect on the differential transmission line mode. The key characteristics are: Continue reading Using SPICE on antenna baluns

Guanella’s 1:1 balun and his explanation – Zcm

Guanella’s 1:1 balun and his explanation gave Guanella’s equivalent circuit and analysis of an example air cored choke of the type shown by Guanella.

The analysis was presented in an LTSPICE model of a ‘test bench’ implementation of the balun, and it showed that on a slightly asymmetric load, common balance was only good in a small region around the choke’s self resonant frequency of 41MHz.

One metric that is useful in indicating the effectiveness of a Guanella 1:1 balun in achieving current balance or reducing common mode current is the choking or common mode impedance Zcm of the stand alone balun.

Modern thinking and experience is that |Zcm| needs to be 1000Ω or higher for effective common mode reduction on many HF wire antennas, and considerably higher for some highly asymmetric antennas.

Zcm of the example air cored solenoid balun

Above is Zcm for the example balun. It is very low at low frequencies and rises to 133+j914Ω at 30MHz. Continue reading Guanella’s 1:1 balun and his explanation – Zcm

Hitachi DB3DL2 UC3SFL repair

The UC3SFL charger for my Hitachi DB3DL2 screwdriver departed this world after just 8 years with a bang, several parts around the main switching transistor deposited as soot inside the cover.

Above is a view of the underside of the charger board, it is very complex, lots of parts, and there are lots of parts on the topside. Ouch, this is going to be expensive.

Yes, it is the replacement power supply is more than half the price of complete tool with two batteries (~$200)… so that is not on.

Modification / fix

The screwdriver’s LED control board has already failed (probably a low grade Chinese electrolytic capacitor) and had to be removed as it would switch on spontaneously and flatten the battery. The hole left vacant by the removed push button provides a convenient exit for a charger cable. Continue reading Hitachi DB3DL2 UC3SFL repair

Guanella’s 1:1 balun and his explanation

It is now 75 years since Guanella’s article “New methods of impedance matching in radio frequency circuits” described the form and operation of what we now commonly call a “Guanella 1:1” balun, but hams being hams also use other terms like “isolator”, “common mode choke”, “current choke” and some insist is it not a balun at all.

In fairness, Guanella did not call the thing a “balun”, but if we accept a very general meaning of balun to be any device intended to facilitate or permit a different state of balance to either side of itself, this is a balun.

Guanella’s description

Above is an extract from (Guanella 1944), and contains an almost complete description of the 1:1 balun. The ideal centre tapped transformers shows are a device for separating the differential and common mode currents so that appropriate elements can be used for those currents. Continue reading Guanella’s 1:1 balun and his explanation

Exploiting your antenna analyser #30

Quality of termination used for calibration

Some of us use a resistor as a load for testing a transmitter or other RF source. In this application they are often rated for quite high power and commonly called a dummy load. In that role, they usually do not need to be of highly accurate impedance, and commercial dummy loads will often be specified to have maximum VSWR in the range 1.1 to 1.5 (Return Loss (RL) from 26 to 14dB) over a specified frequency range.

We also use a known value resistor for measurement purposes, and often relatively low power rating but higher impedance accuracy. They are commonly caused terminations, and will often be specified to have maximum VSWR in the range 1.01 to 1.1 (RL from 46 to 26dB) over a specified frequency range.

Return Loss

It is more logical to discuss this subject in terms of Return Loss rather than VSWR.

Return Loss is defined as the ratio of incident to reflected power at a reference plane of a network. It is expressed in dB as 20*log(Vfwd/Vref). Continue reading Exploiting your antenna analyser #30

EFHW exploration – Part 2: practical examples of EFHW

EFHW exploration – Part 1: basic EFHW explored the basic half wave dipole driven by an integral source as a means of understanding that component of a bigger antenna system.

The EFHW can be deployed in a miriad of topologies, this article goes on to explore three popular practical means of feeding such a dipole.

The models are of the antenna system over average ground, and do not include conductive support structures (eg towers / masts), other conductors (power lines, antennas, conductors on or in buildings). Note that the model results apply to the exact scenarios, and extrapolation to other scenarios may introduce significant error.

End Fed Zepp with current drive

A very old end fed antenna system is the End Fed Zepp. In this example, a half wave dipole at λ/4 height is driven with a λ/4 600Ω vertical feed line driven by a balanced current source (ie an effective current balun).

Above is a plot of the current magnitudes. The currents on the feed line conductor are almost exactly antiphase, and the plot of magnitude shows that they are equal at the bottom but not so at the top. The difference between the currents is the total common mode current, and it is maximum at the top and tapers down to zero at the bottom. Icm at the top is about one third of the current at the middle of the dipole. Continue reading EFHW exploration – Part 2: practical examples of EFHW