The original LCD display was white on blue, but was very difficult to read at some viewing angles, so it had to go. Unfortunately I could not find more displays with that hole pattern, it seems to have been discarded for a newer hole pattern as almost everything I looked at had the same newer patter.

So, the box front needed rework, and there would be visible spare holes… so a dress escutcheon was designed in Freecad and cut on a CNC router.

The escutcheon was designed to be cut from some 3mm black acrylic sheet that was on hand, and it would cover the reworked panel.

One of the nuisances of the ‘modern standard’ pattern LCD1602 is that the edge connector is often very close to a mounting screw hole, too close to comfortably accommodate a piggyback I2C adapter.

Above, relieving the corner with a 8mm diamond cylinder in a Dremel tool allows nut driver access to the mounting screw hole. These are a terrible configuration, as you see the piggyback board is made to require 20mm of space at the side of the module for the i2c plug. In this case, they were bent almost 180° to allow zero clearance at the side.

Above is the reworked rfpm2 with a newer black on yellow LCD which works very well over a wider viewing angle, indoors and outdoors. Some button head screws are on a slow boat from China.

]]>One of the first questions to mind is whether it is likely to deliver the rated power, so let’s review the MOSFET output circuit design from that perspective.

Sellers mostly seem to need to obscure the MOSFET type in their pics, so essentially you buy this with no assurance as to what is supplied, no comeback if the supplied MOSFET is not up to the task. Online experts suggest the MOSFET is probably a MRF9120 (or 2x IRF640 in a 70W build). The amplifier claims 100W from 12-16V DC supply.

Note that this module does not include the necessary output filter which will lose 5-10% of the power from this module.

In this case Carlos, VK1EA, connected a sample output transformer (T2) core from a recently purchased MiniPa100 kit to a EU1KY antenna analyser. The fixture is critically important, it is at my specification.

We also need to know the geometry parameter ΣA/l.

Above, from the measured dimensions of a sample core, ΣA/l=0.003415/m.

The saved S parameter file was processed as described at nanoVNA-H – measure ferrite core permeability described a method for characterising an unknown ferrite material and a complex permeability curve produced.

Above, the results are fairly good and fairly much as expected, but let’s remove the noise by digitising the plots.

Above, the points sampled for the digitised output. Though there is a lot less data in the result, when points are obtained by interpolation, noise is greatly reduced.

The above pic from an eBay advertisement of the 2020 version of the PA would suggest very strongly that there are three turns on the secondary of the output transformer, and a half turn on each drain. Interestingly the 70W versions also appear to use three turns, alarm bells ring!

From all this, we can produce an approximation in Simsmith that captures most of the expected behavior of a practical transformer, including core loss.

Above is the RUSE block schematic used for Core which models the frequency dependent magnetising admittance of the transformer and sets the frequency dependent inductances of the Tfmr element.

(The model assumes that k is independent of frequency which is not strictly correct, but for medium to high µ cores, measurement suggests it is a fairly good assumption.)

More to come…

]]>The radio is an Icom IC-7300. I bypassed the built in tuner, transmitted a tone into my external tuner, adjusted it for SWR=1. I then disconnected the tuner from the radio, and measured the impedance looking into the tuner with a VNA. Surprisingly, (to me anyway) the result was a pretty good 53-j3 Ohms at 14 MHz.

What should we / have expected? It is an interesting case to study.

If:

- the VSWR measurement mentioned was made using the IC-7300 internal VSWR meter;
- if that internal VSWR meter nulled at exactly VSWR(50)=1;
- the second measurement was made by moving the cable connector from the IC-7300 output jack to the VNA; and
- the VNA was OSL calibrated at that jack;

we should expect that the VSWR observed by the VNA was exactly 1.0.

The VNA measurement translated to impedance is reported as 53-j3Ω which implies VSWR(50)=1.09.

What we can say, subject to the numbered conditions listed above, at the power level tested, and allowing for measurement error is that the VSWR meter in the IC-7300 does not seem perfectly accurate, but it is not very far out. Weakness in any of the conditions above might well lead to a conclusion that the test does not show the IC-7300 VSWR has significant error.

What has this to do with optimal matching and the Jacobi maximum power transfer theorem?

Nothing!

It is wrong to interpret this (as the posted seemed to do) as a means of measuring the Thevenin equivalent source impedance of the transmitter. It is an evaluation of the reference (or null) impedance of the IC-7300 internal VSWR meter. An alternative simple test is to connect a high grade 50Ω load and observe the indicated VSWR (which might be power dependent, one of the weakness of the IC-7300) which you might hope to be less than say 1.1… if only you could read it to that resolution (which plays into the accuracy of the reported test).

The IC-7300 VSWR meter is not too easy to read to high resolution, not that it needs to be, but hair splitting tests fail on the uncertainty of such measurements.

None of this addresses whether such a transmitter is well represented by a Thevenin equivalent circuit.

]]>The project was used as the basis for a new project, and during that development work some improvements were made and ported back to RFPM2.

Taking heed of ‘official’ advice that SPIFFS has been deprecated on ESP8266 and that LittleFS was a superior replacement, RFPM2 was migrated to LittleFS.

The main enhancement inherited from ltm is that the choice of configuration file is persistent across restarts. The most recently selected configuration file name is stored in the file system in file /mru.txt, and on restart, that file is read to select a configuration file.

Code remains available on github.

]]>Take a look at the antenna with a VNA and sweep with the Phase function.

Let’s do that!

There are lots of competing firmwares for the nanoVNA, and having tried many and found them wanting, I use the latest firmware from ttrftech, the ‘originator’ of the nanoVNA. So, my comments are in the context of that firmware.

Let’s look at display of the magic phase quantity with a very good load on the nanoVNA, you might think of this load as the ultimate goal of an antenna system.

Above is a screenshot of my nanoVNA where I have selected the ONLY display format labelled phase, and it can be seen that the yellow trace appears to be quite random.

Above is a screenshot of nanoVNA Saver which seems the preferred PC client of the masses, again the same good load is attached. The upper left plot is the ONLY phase plot derived from s11, again it is quite random. Also show are plots of impedance (which is very good), VSWR (which is very good), and Return Loss (which is very good). Return Loss might look noisy, and it is, but it is always greater than 65dB… excellent! The only plot that has NO VALUE in this case is the phase plot!

In fact, when the magnitude of s11 becomes very small, the phase of s11 becomes dominated my measurement noise and it worthless. Yes, the closer you approach the Nirvana of VSWR=1 (ReturnLoss very high, |s11| very low), the less value in the phase of s11.

Let’s look at a sweep of a real antenna, a 5/8λ 144MHz vertical on my car, looking into 4m of RG58 feedline.

On this chart, the easiest curve to interpret for most hams is the VSWR curve (magenta). The markers show its minimum (1.09) and the VSWR=1.5 bandwidth (145.05-150.05MHz)… this is a good antenna from that point of view… but it could be improved by lengthening a little to move the frequency for minimum VSWR down to 147MHz (… but there is no adjustment left).

So, look at the Return Loss blue curve. Return Loss is related to VSWR and you could make exactly the same conclusions. We should accept Return Loss > 15dB.

Look at the s11 phase curve in red. It does not cross the zero phase line (the middle of the chart, in this sweep, it is 44° at minimum VSWR even though it is sometimes less at higher VSWR. Can you make any rational conclusion from the phase curve, and does the fact it is not zero condemn the antenna system?

Look at the R and X curves, green and black. Can you draw any conclusions from them directly? Can you see where the phase of R+jX would be zero? Hint: it is where X is zero… but hey, that doesn’t happen with this antenna system.

After all that information overload, the VSWR curve is the key performance indicator, and I could have used an ordinary VSWR meter to come to the same conclusions pretty much.

Yet another example where the focus on s11 phase is so misguided.

]]>The function t2s is documented in the VNWA help.

t2s is a VNWA built in function intended to solve the so-called s21 series through fixture for impedance measurement of two terminal Zx connected between Port 1 and Port 2.

None of John’s test fixtures were equivalent to the circuit above required for valid t2s transformation.

What if we modify the Simsmith simulated circuit, does series through impedance measurement work?

Above is a minimum change to the simulation circuit to comply with the series through test fixture requirement. The two 25Ω resistors are made 0Ω, and at the other end of the transmission line, both wires are connected together. More on the now redundant elements later.

The figure above is from (Agilent 2009) and it shows the expression for calculation of s21 in the series through configuration. A rearrangement gives \(Z_x=\frac{100}{s_{21}}-100\), which is used in the Simsmith model below.

Above is the simulation showing the calculations used for key values. In element A, Zcm is calculated as 1.413K+j1.04K, and in the G element Plots, Zx is calculated using the algorithm.

Not surprisingly, Zx reconciles with Zcm when the correct test fixture is used.

The simulation circuit above with minimal modification to comply with the series through test requirement now contains redundanct components.

Above, the simulation circuit with the redundant parts removed. It produces exactly the same results, and demonstrates that balun common mode impedance can be measured by connecting the transmission line wires at each end and measuring between the ends. If the transmission line is coax, it is sufficient to measure between the shield ends with the inner conductors left disconnected.

KISS!

The same is true of the simpler s11 reflection method for impedance measurement.

- Agilent. Feb 2009. Impedance Measurement 5989-9887EN.
- Agilent. Jul 2001. Advanced impedance measurement capability of the RF I-V method compared to the network analysis method 5988-0728EN.
- Anaren. May 2005. Measurement Techniques for Baluns.
- Skelton, R. Nov 2010. Measuring HF balun performance in QEX Nov 2010.

Above is the subject balun in fixture.

John’s schematic shows the balun as coupled coils, but that does not capture the transmission line transformation that occurs in the actual device. Again the test fixture is used without explanation.

To implement a transmission line model of the balun, we need to capture both its transmission line behavior to differential currents, and its choke behavior to common mode current. To do that, two pairs of coupled coils are added at each end of the transmission line to divert the common mode current via the choke elements.

So, now we can reasonably accurately model the transmission line effect and the choke effects.

Above is the revised Simsmith model that implements the schematic above, and compares the model with measured for that balun configuration and test fixture. Again the value of n was tweaked to calibrate the model, and Cse adjusted for good high end tracking.

Above is my reinterpretation of his measurement data.

Looking at the marker values on my chart, they indicate Zx=5640+j4321Ω @ 14MHz, much higher than the expected common mode impedance of the choke. They are of course nonsense, the t2s function is not a valid transformation for John’s test fixture.

The test fixture used seems inspired by the theme ‘more complicated is naturally better’. KN5L’s model is flawed.

- Agilent. Feb 2009. Impedance Measurement 5989-9887EN.
- Agilent. Jul 2001. Advanced impedance measurement capability of the RF I-V method compared to the network analysis method 5988-0728EN.
- Anaren. May 2005. Measurement Techniques for Baluns.
- Skelton, R. Nov 2010. Measuring HF balun performance in QEX Nov 2010.

(Anaren 2005) explains a method of finding balun CMRR. Anaren gives a definition of CMRR:

Common Mode Rejection Ratio is defined and the ratio between the differential mode insertion loss/gain versus the common mode signal loss or gain.

Note that in a passive system, CMRR in dB will usually be positive, and the larger the better.

Anaren does not mention applying the CMRR statistic to antenna systems. I have commented elsewhere on the lack of utility of CMRR in analysing common antenna systems.

John, KN5L, has published his own solution to balun characterisation in some online forums.

Let’s look at his example with a 7t RG174 winding on a FT140-43 core in his recommended test fixture. He does not give a schematic of the test fixture, but it can be gleaned from his pic.

At the left is VNA Port 1 connection, the coax connects shield to the coax connector outer, and coax inner to connector inner. At the right hand end of the coax, the shield connects to the connector inner via a series 25Ω resistor, and the coax inner connects to the connector inner via another series 25Ω resistor, both coax connector outers are connected to the PCB copper plane, and the right hand coax connector connects to the VNA Port 2. He gives no explanation of why such a test fixture was chosen.

Above is his published VNA sweep. Analysing his published .s2p file reveals that the curves labelled “RG174 CMRR” is in fact |s21|… so he has implied his own meaning for CMRR, he does not give a clear definition other than this implication. The negative values of “RG174 CMRR” sound a warning.

John publishes a Simsmith model comparing a theoretical model of the balun in his test fixture with measurement. His Simsmith model of the balun in fixture is flawed, so I will use my own.

In this case, I have tweaked the number of turns a little to get a closer fit between model and measurement, ferrite has quite wide tolerance and the model is simple so we should not expect exact reconciliation.

Also calculated is the expected balun common mode impedance Zcm, in this case 1063+j617Ω @ 14MHz.

Above is my chart of his published measurement file. The curve “s21 dB” is simply |s21| from Port 1 to Port 2 through his test fixture, it is not in accord with Anaren’s definition of CMRR as he labels his plots.

The curves “t2s(s21) real Z” and “t2s(s21) real Z” mimic a calculation John gives on some other examples. The function t2s is documented in the VNWA help.

t2s is a VNWA built in function intended to solve the so-called s21 series through fixture for impedance measurement of two terminal Zx connected between Port 1 and Port 2.

Looking at the marker values on my chart, they indicate Zx=5554+j4366Ω @ 14MHz. Is that the true common mode impedance? Hhe seems to be saying that:

ALL previous NEC Balun CM current models using a single inductor to

simulate two flux coupled inductors, of the same value, are flawed.

Well, applying the t2s function to data from a different test fixture circuit is invalid, the results are invalid, conclusions drawn from it are invalid.

NEC is quite capable of modelling the common mode current path separately to a TL element (which models only the differential mode), the appropriate value to load the common mode conductor path with is the calculated Zcm (which is frequency dependent).

The test fixture used seems inspired by the theme ‘more complicated is naturally better’.

- Agilent. Feb 2009. Impedance Measurement 5989-9887EN.
- Agilent. Jul 2001. Advanced impedance measurement capability of the RF I-V method compared to the network analysis method 5988-0728EN.
- Anaren. May 2005. Measurement Techniques for Baluns.
- Skelton, R. Nov 2010. Measuring HF balun performance in QEX Nov 2010.

Reflection Bridge and Return Loss Bridge are somewhat synonymous, in practice to measure Return Loss one is interested in the magnitude of the response, and to measure the complex reflection coefficient or s11, both magnitude and phase are of interest.

Above is Oristopo’s graph.

We can create a model of a Return Loss Bridge or Reflection Bridge in Simsmith and plot its response for swept Zu.

Above is the Simsmith model with plot for R swept from 1-2500Ω and X=0Ω (for simplicity). When plotted on a log frequency scale, the |s11| response is symmetric, and the markers at 50/10 and 50*10 both produce |s11|=-1.73dB.

Note that this simulated bridge complies with ALL the requirements for correct response:

- source impedance is 50+j0Ω by virtue of the G element definition;
- the three known elements of the bridge are specified as 50(+j0)Ω;
- the bridge detector load impedance is 50+j0Ω by virtue of the definition of element L.

For convenience, the source power is defined as 16W so that it produces 1W or 0dBW at the detector when Zu=0 (or ∞).

The calculated |s11| (or -ReturnLoss) can be calculated easily to verify these two cases using a calculator, or good online calculator. For example using Calculate VSWR and Return Loss from Zload (or Yload or S11) and Zo.

From a point of digitising the s11 response, the challenge is as great for 5Ω as it is for 500Ω. Very low and very high impedances are sensitive to different aspects of the fixture, so it is easy to make a fixture that compromises high or low impedances.

When |s11| becomes relatively large (ie approaching 1, or 0dB) as it does for measurement of very high and very low impedances, the ADC resolution becomes an issue, internal noise of the instrument becomes significant, and accurate phase measurement is more difficult, and as a result, measurement accuracy is compromised.

Several recent articles have used measurements of transmission line sections with SC and OC terminations.

Above is an example where at HF, |s11| >-0.05dB, which is the magnitude of |s11| with a load of 17370+j0Ω, or 0.1439Ω. Sure, there is some noise, but the measurements are usable for the purpose at hand.

One wonders if some online experts have condemned high impedance measurements as grossly inaccurate based on their own experience, perhaps with flawed fixtures, maybe they are just quoting another online expert they have read.

Generalised assertions by online experts that VNAs cannot accurately measure impedance above a few hundred ohms are not borne out by careful measurement experience of known DUTs in appropriate fixtures… or they have unreal expectations about the accuracy required for common analyses.

]]>Oristopo gives a diagram and explanation.

Above is his diagram. He gives an expression that he states applies when R1=R3=R4=Rm: im = sqrt(Vf*(Rm – R2)/(12*Rm + 4*R2)).

This is deeply flawed, if R2>Rm the expression results in the square root of a -ve number… which might be acceptable in a complex number scenario, but this is a DC circuit.

Nevertheless, let us calculate the current with R2=0 and R2=5 while R1=R3=R4=Rm=50.

- R2=0: Im=0.28867513459481287;
- R2=5: Im=0.26940795304016235.

We can calculate \(ReturnLoss=20 log10 \frac{0.28867513459481287}{0.26940795304016235}=0.6 \;dB\).

Wrong, the ReturnLoss of a 5Ω load on a 50Ω Return Loss Bridge should be 1.7dB.

So, the circuit / expression does not have the response of a Return Loss Bridge.

That is understandable, the schematic is not that of a Return Loss Bridge. If a Return Loss bridge uses R1=R3=R4=Rm, then its source MUST also have a source impedance Rs where Rs=R1=R3=R4=Rm for an accurate Return Loss response.

Analysis based on the schematic above with Zs=0 is not representative of the Return Loss Bridge used in accurate instruments, and conclusions are not soundly based.

The Return Loss Bridge is a deceptively simple thing… it does take careful attention to all details to obtain accurate results.

In the more general sense of a VNA, the reflection bridge must respond proportionally to the complex reflection coefficient.

Oristopo’s graph would imply that the nanoVNA can measure down to zero ohms but not above a few hundred ohms. The simple fact is that a reflection bridge calibrated for 50Ω returns the same magnitude voltage for 2.5+j0Ω load as for 1000+j0Ω, just the phase is opposite… so if measurement noise is a problem for one, it is likely to be much the same for the other.

Generalised assertions by online experts that VNAs cannot accurately measure impedance above a few hundred ohms are not borne out by careful measurement experience of known DUTs in appropriate fixtures… or they have unreal expectations about the accuracy required for common analyses.

]]>