Above is an analysis of KN5L’s published measurements of a 19.93m test section of Seminole 1320 (nominal 300Ω windowed ladder line, 0.812mm (#20) 7 strand copper). The line was purchased around 2015. The plot has:
It can be seen that:
Above is an analysis of KN5L’s published measurements of a 14.54m test section of Seminole 1321 (nominal 300Ω windowed ladder line, 1.024mm (#18) 19 strand copper clad steel). The line was purchased around 2015. The plot has:
It can be seen that:
The measured MLL at 1MHz is 0.12dB/m.
Above is a model of current distribution in a 1.024mm (#18) round CCS conductor with cladding equal to that of the component 0.255mm (#30) strands 30% IACS (17.8µm) as an approximation of the #30×19 conductor. That model suggests that the MLL is 0.11dB/m. We can fully expect that the loss of a stranded conductor will be a little higher, so measurement and prediction reconcile reasonably well.
Loss of ladder line: copper vs CCS (DXE-LL300-1C) – revised for 25/07/2018 datasheet was a revision of an earlier article based on an updated datasheet from DXE. I noted that the specification data had artifacts that one would not expect of such a line, and I questioned whether the datasheet was credible.
John, KN5L, recently purchased, measured and published measurements of a 10.06m (33′) section of new DXE-LL300-1C which provide an independent dataset that might cast some light on the matter.
The chart above plots:
The theoretical line is based on well developed skin effect and \(MLL \propto \sqrt f\), resulting in a straight line on the log-log graph.
KN5L’s 100 point measurement dataset is for the most part smooth and quite credible, though it shows departure from ideal homogenous conductors with well developed skin effect… and for good reason, these are not homogenous conductors and skin effect is only developed at higher frequencies.
At 60MHz, KN5L’s measured MLL is a little worse than theoretical, quite probably due to the fact that these are 19 strand conductors, and the cladding thickness may be just too little to deliver copper like performance even at 60MHz. At lower frequencies, MLL is better, but a good deal worse than the theoretical MLL for copper conductors.
Whilst the MLL might seem small, these types of line are commonly used in scenarios with high VSWR. Let’s calculate the loss under mismatch of a scenario used for some recent articles.
The scenario then is the very popular 132′ multi band dipole:
We will consider the system balanced and only deal with differential currents.
Taking the MLL of the LL300 as 0.018dB/m, the calculated loss under mismatch is 2.3dB. It is not huge, but any assumption that the loss in open wire line is insignificant is wrong.
Lets evaluate the loss using a home made open wire line of 2mm copper conductors spaced 150mm. The calculated loss under mismatch is 0.114dB, a lot better than the previous case.
]]>Above is a pic of the Rose Clip as a line insulator / support. The clips are pretty flimsy but pretty cheap. They click onto the 1.6mm diameter wire reasonably firmly.
Above, a small dob of hot melt glue is used to fix the clip at one end only so that it does not move along the line. The other side is clicked on but free to move to accommodate twists and turns. The clips will probably be needed at intervals of 0.5m or perhaps less in some situations.
The short test section has been in the weather for two years, but has not been subjected to a real trial as a long section of line in all weather.
Above is a calculation of expected performance. The velocity factor is a guess based on experience.
Essentially, though it is very low cost, it achieves Zo around 480Ω with matched line loss around 0.23dB/100m @ 3.5MHz. The clips cost about $6/100 and the wire about $0.07/m, so all up $0.26/m.
Above is a mockup of a dipole leg terminated on an insulator, and the feed line connected using the 3mm stainless steel tap connector. In use, the tap connector will be filled with marine grease or aluminium jointing compound to exclude oxygen and water.
Above is a 2mm stainless connector. The 1.6mm wire has to be bent to follow the path through this smaller connector body. Again it needs to be filled with grease to exclude oxygen and water. These connectors are available on eBay at low cost.
]]>The original scenario then is the very popular 132′ multi band dipole:
We will consider the system balanced and only deal with differential currents, and matched line loss is based on measurement of a specific sample of line (RG6/U with CCS centre conductor at HF).
This article will calculate the same scenario with three feed line variants:
The loss under mismatch depends not only on the transmission line characteristics and length, but also on the load and the current and voltage distribution.
Above the 150Ω twin line with same CCS conductors as the RG6 has loss almost identical to the synthesised twin shielded in the original article. Almost all of the resistance in the coax is in the CCS centre conductor, so I assume that the loss in the twin CCS is approximately equal to that of the synthesised twin. Dielectric loss is less than 1% and can be ignored.
Above the 600Ω twin line with same CCS conductors as the RG6 (ie the spacing is increased to increase Zo). Almost all of the resistance in the coax is in the CCS centre conductor, so I simply assume that the 600Ω twin line with same CCS conductors has 150/600 times the matched line loss. The loss is considerably lower at 0.354dB in this scenario, due to the higher Zo.
Above the 600Ω twin line with 2mm HDC. The loss is considerably lower again at 0.061dB in this scenario, due to the higher Zo.
Note that in all these cases, the load impedance and length of the line form an important part of the evaluation scenario.
So, we can identify that two factors result in the quite poor performance of the synthesised shielded twin:
Improving both of these factors in the third scenario reduces loss under this mismatch scenario by a factor of 50.
]]>These were very popular at one time, but good voltage baluns achieve good current balance ONLY on very symmetric loads and so are not well suited to most wire antennas.
Above is a pic of the balun with load on test. It is not the greatest test fixture, but good enough to evaluate this balun over HF.
Mine has survived, but many users report the moulding cracking and rusted / loose terminal screws, and signs of internal cracks in the ferrite ring.
InsertionVSWR is often an important parameter of nominally 1:1 baluns. So, let’s measure the balun’s InsertionVSWR by connecting it to Port 0 (Ch0) and connecting a good 50+j0Ω load to the output terminals.
Above is a sweep from 1-41MHz, Insertion VSWR looks pretty good above about 7MHz and up to about 20MHz.
Let’s drill down on the low frequency performance.
Above is a Smith chart view of the sweep from 1-5MHz. If you are familiar with the Smith chart, you will recognise that the curve almost follows a circle of constant G (not drawn on this Smith chart unfortunately). That suggests that Yin is approximately 1/50+jB where B is frequency dependent.
nanoVNA MOD does not have an admittance chart (more’s the pity) but it does have a hammy substitute, the “parallel RLC” chart though it is actually Rp||Xp. Let’s sweep 1-5MHz to focus on the low frequency InsertionVSWR problem.
Above, the Rp||Xp presentation. Note that the Xp (blue line) is fairly straight and if you project it to frequency=0, Xp will be approximately 0 so \(Xp \propto f\). Recall that the reactance of an inductance X=2πfL, so Xp looks like it may be due to a constant parallel inductance, and the equivalent parallel inductance can be calculated. It can be, but no need as nanoVNA MOD conveniently displays the value in the cursor data, 11.9µH in this case. Note also that the value of Rp is approximately 50Ω independent of frequency.
The poor low frequency InsertionVSWR is due to the low equivalent parallel inductance of 11.9µH at low frequencies, the magnetising inductance as it happens.
So, what should it be?
Well that depends on how we might specify performance. If we wanted the balun to have an InsertionVSWR of less than 1.1 from 3.5MHz, then Xp needs to be greater than \(Xp>10Zo=500\Omega\) and therefore magnetising inductance \(Lp>\frac{10 Zo}{2 \pi f}=23µH\).
Increasing the magnetising inductance will typically degrade the high frequency performance, so finding a good design is a compromise between these and other factors.
If we look more widely at the Rp||Xp response, we see a self resonance around 22MHz, and above that, progressively a lower and lower shunt Xc. So, just as low equivalent shunt Xl degraded low end performance, low equivalent shunt Xc degrades high end performance which is the main contribution to increasing InsertionVSWR above 22MHz.
So as voltage baluns go, this has moderately good InsertionVSWR from 7-20MHz, but is a bit shabby above and below that range.
The article has demonstrated how simple measurements made with the nanoVNA (or any other capable VNA or antenna analyser) can be used to evaluate not only the InsertionVSWR, but provide a likely explanation for its behaviour. Insufficient magnetising impedance is a common design flaw. You could use this approach to guide design of a DIY voltage balun.
Further reading: Voltage symmetry of practical Ruthroff 1:1 baluns discusses voltage symmetry of the BL-50A.
]]>This article revises Loss of ladder line: copper vs CCS (DXE-LL300-1C) for revised published datasheet MLL figures with internal PDF date of 25/07/2018.
Let’s start by assuming that the new offered data is credible, let’s take it at face value.
The line is described as 19 strand #18 (1mm) CCS and the line has velocity factor (vf) 0.88 and Zo of 272Ω.
Let us calculate using TWLLC the loss at 2MHz of a similar line but using pure solid copper conductor with same conductor diameter, vf and Zo. We will assume dielectric loss is negligible at 2MHz
Parameters | |
Conductivity | 5.800e+7 S/m |
Rel permeability | 1.000 |
Diameter | 0.00100 m |
Spacing | 0.00650 m |
Velocity factor | 0.880 |
Loss tangent | 0.000e+0 |
Frequency | 2.000 MHz |
Twist rate | 0 t/m |
Length | 30.480 m |
Results | |
Zo | 272.69-j2.59 Ω |
Velocity Factor | 0.8800 |
Length | 83.18 °, 0.231 λ, 30.4800 m, 1.155e+5 ps |
Line Loss (matched) | 0.121 dB |
Spacing has been adjusted to obtain Zo.
At 2MHz MLL of a copper line is 0.121dB for 30.48m (100′) as against 0.32dB measured for the stranded CCS line.
At 50MHz MLL of a copper line is 0.641dB for 30.48m (100′) as against 0.89dB measured for the stranded CCS line.
If the measurement data was valid and correct, the difference would almost certainly attributable to CCS and stranding. The copper cladding on the very thin strands is way less than skin depth at lower frequencies, effective RF resistance is higher than that of a solid copper conductor.
You might regard that the difference is tenths of a dB and insignificant, but this line is almost always used at high VSWR and the difference between the two lines is likely to be significant.
If we take the measured data and fit a model that matched line loss is per unit length of line (m) is:
\(MLL=(k_1 \sqrt f + k_2 f)l\)
Where | Loss = | loss per unit length |
f = | frequency | |
k1 = | constant | |
k2 = | constant | |
l = | length |
Such a model is usually a good fit for practical transmission lines where skin effect is well developed, and dielectric loss is proportional to frequency. A solution for k1 and k2 for least squares error has been found for the DXE published data.
Above is a plot of the measured data and the model.
The measured data curve exhibits some form of oscillation about some possibly smoother curve. The oscillation is unexpected and ought prompt review of the measurement setup to see that there is not some other effect being captured, eg unbalanced drive exciting common mode resonances.
Nevertheless, it we treat the data as correct, the issue that arises is that the value for k2 is significantly negative, and we ought to expect it is positive and smaller than k2 at these frequencies.
We might expect and excuse some obvious departure from the model at frequencies below 5MHz due to the copper clad steel conductors.
So, the extent of oscillation and higher frequencies and poor fit to the model raises some questions about the validity of the measurement data.
]]>These were very popular at one time, but good voltage baluns achieve good current balance ONLY on very symmetric loads and so are not well suited to most wire antennas.
Above is W2AU’s illustration of the internals.
Mine barely saw service before it became obvious that it had an intermittent connection to the inner pin of the coax connector. That turned out to be a poor soldered joint, a problem that is apparently quite common and perhaps the result of not properly removing the wire enamel before soldering.
Having cut the enclosure to get at the innards and fix it (they were not intended to be repaired), I rebuilt it in a similar enclosure made from plumbing PVC pipe and caps, and took the opportunity to fit some different output terminals and an N type coax connector.
Above is the rebuilt balun which since that day has been reserved for test kit for evaluating the performance of a voltage balun in some scenario or another.
My rework did not attempt to duplicate the spark gap arrangement of the top terminals. It is doubtful that it is effective protection of an attached receiver.
InsertionVSWR is often an important parameter of nominally 1:1 baluns. So, let’s measure the balun’s InsertionVSWR by connecting it to Port 0 (Ch0) and connecting a good 50+j0Ω load to the output terminals.
Above is a sweep from 1-41MHz, Insertion VSWR looks pretty good above about 7MHz.
Let’s drill down on the low frequency performance.
Above is a Smith chart view of the sweep from 1-5MHz. If you are familiar with the Smith chart, you will recognise that the curve almost follows a circle of constant G (not drawn on this Smith chart unfortunately). That suggests that Yin is approximately 1/50+jB where B is frequency dependent.
nanoVNA MOD does not have an admittance chart (more’s the pity) but it does have a hammy substitute, the “parallel RLC” chart though it is actually Rp||Xp. Let’s sweep 1-5MHz to focus on the low frequency InsertionVSWR problem.
Above, the Rp||Xp presentation. Note that the Xp (blue line) is fairly straight and if you project it to frequency=0, Xp will be approximately 0 so \(Xp \propto f\). Recall that the reactance of an inductance X=2πfL, so Xp looks like it may be due to a constant parallel inductance, and the equivalent parallel inductance can be calculated. It can be, but no need as nanoVNA MOD conveniently displays the value in the cursor data, 12.9µH in this case. Note also that the value of Rp is approximately 50Ω independent of frequency.
The poor low frequency InsertionVSWR is due to the low equivalent parallel inductance of 12.9µH, the magnetising inductance as it happens.
So, what should it be?
Well that depends on how we might specify performance. If we wanted the balun to have an InsertionVSWR of less than 1.1 from 3.5MHz, then Xp needs to be greater than \(Xp>10Zo=500\Omega\) and therefore magnetising inductance \(Lp>\frac{10 Zo}{2 \pi f}=23µH\).
Increasing the magnetising inductance will typically degrade the high frequency performance, so finding a good design is a compromise between these and other factors.
So as voltage baluns go, this has quite good InsertionVSWR above 7MHz, but is a bit shabby below that.
The article has demonstrated how simple measurements made with the nanoVNA (or any other capable VNA or antenna analyser) can be used to evaluate not only the InsertionVSWR, but provide a likely explanation for its behaviour. Insufficient magnetising impedance is a common design flaw. You could use this approach to guide design of a DIY voltage balun.
]]>In respect of the first part, inductance \(L=\frac{\phi(i)}{i}\) so if the windings are equal, half the total current flows in each winding and each contributes flux due to i/2, total current is i, total flux is twice that due to i/2, so the inductance of the parallel equal windings is the same as if i flowed in a single winding, ie L of the combination is the same as the inductance of each of the equal windings alone.
In the second case, if there is zero flux leakage, it is true that the inductance of the combination of series opposed equal windings is zero. In the case of the cores used for common mode chokes flux leakage varies from less than 1% to over 50%… so the general statement is a bit naive in this application.
Both of these are very easily demonstrated by simple experiment. In fact a measurement of L with series aiding and series opposing connection is a classic way to find the flux coupling factor.
A few years after the ‘information’ was posted, it has not been questioned much less called out as wrong. So given that the understanding of inductance seems lacking… what credibility does the article have when it includes this?
]]>fenceas explained in the following text by one poster.
With a current balun or CM choke, it is the reactance (inductance) that is mostly responsible for the balun action. In the case of the choke balun, beads installed along the coax at the feed with 31 or 43 material, they form a reflective ‘filter’. There is some absorption, but most of the action is due to reflection from the inductive reactance they form installed on a conductor. As such, they form a high-Z isolation point between the feeder and the antenna center, assuming they are installed at the feedpoint of the doublet. In the case of the CM choke, the common mode currents are reflected by the inductive reactance of the windings as with the current balun and the balance of current between the two conductors is forced through induced opposing magnetic currents within the cone. This is the reason I prefer the CM choke for the purpose. In either case, the common mode current is reflected to a large extent by the inductive reactance back where it originated. Installation of a balun at the feedpoint of a doublet does not make the CM currents go away, it just establishes a ‘fence’ for those currents between non-antenna associated currents (on the outside of the feedline) and the radiating structure.
Let us explore some NEC models with three ‘devices’ to attempt to confine current to the lower conductor:
The first is at 10MHz a vertical conductor over a perfectly conducting earth, and space 0.1m above it, another vertical conductor.
Above is the current distribution showing phase and amplitude, the gap is at one third the height. It is not totally clear from the 2D rendering of a 3D characteristic, but the phase in the upper two thirds is opposite to the phase in the lower third, and this is by virtue of the lengths which are approximately a quarter and half wavelength.
So, even where there is a substantial gap, the upper conductor is coupled to the lower conductor and the current is not confined to the lower conductor.
The gap in the above model was filled with a conductor loaded with Z=0+j1000Ω.
Above is the current distribution showing phase and amplitude, the gap is at one third the height.
A view from the zenith of current magnitude and phase plot shows the phase in the upper part is about 120° out of phase with the lower third.
So, even where there is a large pure inductive reactance, the upper conductor is coupled to the lower conductor and the current is not confined to the lower conductor.
The gap in the first model was filled with a conductor loaded with Z=1000+j0Ω.
Above is the current distribution showing phase and amplitude, the gap is at one third the height. In this case the phase in the upper two thirds is close to opposite that in the bottom third.
So, even where there is a large pure resistance, the upper conductor is coupled to the lower conductor and the current is not confined to the lower conductor.
None of these simple structures isolate the current to the lower element only, there is significant current in the upper element, indeed greater current moment in the latter two cases by virtue of its length.
These experiments suggest that behavior is not simply as set out in the quote.
]]>Above is the active whip antenna. Not optimal mounting, but as you can see from the clamps, a temporary mount but one that does not confuse results with feed line common mode contribution.
Above is the remote electronics, the RPi 3B+ and RDL SDR dongle, and underneath the power supply for the active monopole.
The initial trial was on 7MHz, and was a total failure due to extreme level of RFI from the RPi itself. The two RPi power supplies tested were noise tested on a dummy load and were OK, the noise comes from the digital signals on the RPi board. A Kenwood R-5000 receiver was connected to the active antenna, and noise floor was relatively low until the RPi was plugged in.
Above is the SdrSharp screen, extreme noise level and no signals could be heard (though a local transmitter on low power verified that the receiver was working).
The emissions from the RPi, were so high that it is really unsuitable for this purpose, it would be very difficult to reduce emissions by the needed more than 40dB.
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