Above is the transformer with 100pF compensation capacitor across the input, and two resistors to make up a 3300Ω load in combination with the VNA port.
The transformer is an autotransformer of 16t tapped at 2t, so the nominal turns ratio is 1:8, or impedance 1:64.
Looking into the 2t tap with the transformer loaded with 3250+50=3300Ω, Luis measured s11, s21 parameters from 1-45MHz with a nanoVNA. (Nominally this should be a 3200Ω load, but the error is very small, VSWR equivalent of 1.03.)
Above is my analysis of this data for InsertionVSWR and ReturnLoss. It gets a bit shabby above 15MHz.
Above is a plot of InsertionLoss and Loss (or TransmissionLoss). See On Insertion Loss for explanation of the terms.
Compensation of the transformer with 100pF in shunt with the input improves the broadband response.
Above is InsertionVSWR and ReturnLoss. It is not too bad over all of HF (3-30MHz).
Above is a plot of InsertionLoss and Loss (or TransmissionLoss). See On Insertion Loss for explanation of the terms.
Almost all of the Loss is in core loss, and it gives us a good indicator of core heating. The worst case is at 30MHz where the Loss is 0.7dB, so about 15% of input power is converted to heat.
The tests here were using a dummy load on the transformer, and that did allow confirmation of the design.
Real end fed antennas operated harmonically do not present a constant impedance, not even in harmonically related bands. Note that the resonances do not necessarily line up harmonically, there is commonly some enharmonic effect.
Being a more efficient design that some, it might result is a wider VSWR excursion that those others as transformer loss can serve to mask the variations in the radiator itself.
Hats off to Luis, CT2FZI for his work in building and measuring the transformer, and to have the flexibility to consider more than FTxxx-43 cores.
]]>Above is the model topology. D1 is a daemon block which essentially, calculates key values for the other blocks based on exposed parameters and the named ferrite material complex permeability data file. The prototype used a Fair-rite 2643625002 (#43) core.
D1 code:
//Misc //Updates Tfmr, CoreLoss, and Cse. $data=file[]; // core mu aol; np; ratio; cse; cores; k; $u1=$data.R; $u2=$data.X; //u1=$u1; //u2=$u2; $ns=np*ratio; Ym=(2*Pi*G.MHz*1e6*(4*Pi*1e-7*$u1*aol*cores*1e9)*np^2*1e-9*(1j+$u2/$u1))^-1; Cse.F=cse; CoreLoss.ohms=1/Ym.R; $l1=1/(2*Pi*G.MHz*1e6*-Ym.I); $l2=$l1*(($ns-np)/np)^2; $lm=k*($l1*$l2)^0.5; Tfmr.L1_=$l1+$lm; Tfmr.L2_=$l2+$lm; Tfmr.L3_=-$lm;
L is the load block.
Cse models the self resonance of the transformer at lowish frequencies, Cse is an equivalent shunt capacitance.
Tfmr is a coupled coils model (above) of a (nearly) lossless autotransformer (core loss will come later). Wire loss is usually insignificant in these type of transformers and is ignored. (It seems that Simsmith will not simulate a pure inductance, in this cases the loss of the ‘lossless transformer is of the order of 5e-6dB, so satisfactory.) It is possible to simplify declaration of this component by using Simsmith’s coupled coils in a RUSE block, I have chosen to take full control and solve the mutual inductance effects explicitly.
(The model assumes that k is independent of frequency which is not strictly correct, but for medium to high µ cores, measurement suggests it is a fairly good assumption.)
Coreloss brings the ferrite core loss to book.
Ccomp models a compensation capacitor used to improve broadband InsertionVSWR.
The G block provides the source and plot definitions.
Plots code:
//Plots //check lossless Tfmr behavior //tfmrloss_dB=10*Log10(Tfmr.P/(Tfmr.P-Tfmr.p)); //Plot("TfmrLoss",tfmrloss_dB,"LossdB",y1); coreloss_dB=10*Log10(CoreLoss.P/(CoreLoss.P-CoreLoss.p)); loss_dB=10*Log10(G.P/L.P); mismatchloss_dB=10*Log10(1/G.P); Plot("CoreLoss",coreloss_dB,"LossdB",y1); Plot("Loss",loss_dB,"LossdB",y1);;
The model was calibrated to measurement of the prototype, and the fit is quite good given tolerances on components.
The model allows convenient interactive sensitivity analysis where parameters can be dialed up and down with the mouse wheel and the response changes observed.
At Ultrafire XML-T6 LED torch – a fix for the dysfunctional mode memory ‘feature’ I gave a fix for that revision of the electronics, and updated it with description of a later fixed production model.
Years later, I bought two more of these due to switch failures on the originals… and guess what, the flash on power on returns.
Let’s pull them apart.
They have a new revision / version of the LED driver PCB, and it has provision for a resistor in parallel with the capacitor, but the resistor pads are not populated.
Above, the LED driver board with a 100k resistor added, it is the far component. This was an 0805 part that was on hand, but ideally should be a 0603.
Above, the second torch could not be saved, the driver PCB was cracked, a result of heavy handed forcing the PCB into its recess when not properly aligned.
This model is a little different construction to the first ones I wrote about. To dismantle it:
To finish, add the resistor, clean everything, wipe a little silicon grease inside the front barrel, slide the LED housing into it and screw the main barrel onto it. Tighten the LED housing, install the battery, lubricate the rear thread and o-ring with silicon grease, and test it all works. Then clean the LED, reflector and lens and reinstall the lens and retaining ring. All done.
]]>I bought this after seeing several recommendations on a nanoVNA forum.
Above is the factory pic of the SMA-8.
The one I purchased cost about $40 including post from an eBay store.
In the event, the thing arrived after nearly three weeks… so that was quick.
When the mechanism was operated by hand, it was obviously asymmetric.
But it came with a test certificate!
So I will measure it.
Above, force from the left peaks at 6.0N, @ 0.16m radius: 0.96Nm torque.
Above, force from the right peaks at 9.4N, @ 0.16m radius: 1.5Nm torque.
The thing is unusable and unrepairable.
I trust my measurements before the obviously phony Chinese test certificate.
This was purchased on eBay with the hope that it would be better quality than the last, and if it wasn’t there was better prospect of a refund. A full refund was obtained after making a compelling case.
I am not saying all MXITA product is low quality, but this item was such poor quality as to be consigned to the bin and if you trust MXITA as I did, you might do the same.
]]>Let’s demonstrate the measurement of Rin of an o/c resonant section around the 160m band, which we will then use to calculate MLL.
Above, the AA-600 connected to the cable using a F(F)-N(M) adapter, the cable is 305m in length and the far end is open circuit.
So lets scan around 1800kHz looking for a convenient resonance.
Ok, there is a resonance just a bit above 1800kHz. Let’s move the cursor using the freq+ arrow key, press the range- key a few times to narrow the scan and press to rescan.
Ok, closer now, but overshot a bit. Let’s move the cursor again and switch to All mode.
Above is All mode, X is just greater than zero, so we will shift freq down using the freq- arrow key watching X for the zero crossing.
Above, this is close enough. We can read Rin as 19.0Ω.
Above, calculating MLL from the input data we have 0.0074dB/m or 0.74dB/100m.
This whole procedure takes less than a minute using the stand alone instrument… in part due to the very effective human interface on the AA-600, you might think the designer knew how the instruments would get used, something you cannot say for a lot of analysers and VNAs (which are now the fashion).
The technique is particularly useful if the far end of the cable is not accessible, eg it is sometimes hidden in the interior of the drum.
(Ikin 2016) proposes a different method of measuring noise figure NF.
Therefore, the LNA noise figure can be derived by measuring the noise with the LNA input terminated with a resistor equal to its input impedance. Then with the measurement repeated with the resistor removed, so that the LNA input is terminated by its own Dynamic Impedance. The difference in the noise ref. the above measurements will give a figure in dB which is equal to the noise reduction of the LNA verses thermal noise at 290K. Converting the dB difference into an attenuation power ratio then multiplying this by 290K gives the LNA Noise Temperature. Then using the Noise Temperature to dB conversion table yields the LNA Noise Figure. See Table 1.
The explanation is not very clear to me, and there is no mathematical proof of the technique offered… so a bit unsatisfying… but it is oft cited in ham online discussions.
I have taken the liberty to extend Ikin’s Table 1 to include some more values of column 1 for comparison with a more conventional Y factor test of a receiver’s noise figure.
Above is the extended table. The formulas in all cells of a column are the same, the highlighted row is for later reference.
A test setup was arranged to measure the noise output power of an IC-7300 receiver which has a sensitivity specification that hints should have a NF≅5.4dB. The relative noise output power for four conditions was recorded in the table below.
Ikin’s method calls for calculating the third minus second rows, -0.17dB, and looking it up in his table. In my extended table LnaNoiseDifference=-0.17dB corresponds to NF=3.10dB.
We can find the NF using the conventional Y factor method from the values in the third and fourth rows.
The result is NF=5.14dB (quite close to the expected value based on sensistivity specification).
Ikin’s so called dynamic impedance method gave quite a different result in this case, 3.10 vs 5.14dB, quite a large discrepancy.
The chart above shows the relative level of the four measurements. The value of the last two is that they can be used to determine the NF using the well established theory explained at AN 57-1.
The values in the first columns are dependent on the internal implementation of the amplifier, and cannot reliable infer NF.
Let’s review of the concepts of noise figure, equivalent noise temperature and measurement.
Firstly let’s consider the nature of noise. The noise we are discussing is dominated by thermal noise, the noise due to random thermal agitation of charge carriers in conductors. Johnson noise (as it is known) has a uniform spectral power density, ie a uniform power/bandwidth. The maximum thermal noise power density available from a resistor at temperature T is given by \(NPD=k_B T\) where Boltzman’s constant k_{B}=1.38064852e-23 (and of course the load must be matched to obtain that maximum noise power density). Temperature is absolute temperature, it is measured in Kelvins and 0°C≅273K.
Noise Figure NF by definition is the reduction in S/N ratio (in dB) across a system component. So, we can write \(NF=10 log \frac{S_{in}}{N_{in}}- 10 log \frac{S_{out}}{N_{out}}\).
One of the many methods of characterising the internal noise contribution of an amplifier is to treat it as noiseless and derive an equivalent temperature of a matched input resistor that delivers equivalent noise, this temperature is known as the equivalent noise temperature Te of the amplifier.
So for example, if we were to place a 50Ω resistor on the input of a nominally 50Ω input amplifier, and raised its temperature from 0K to the point T where the noise output power of the amplifier doubled, would could infer that the internal noise of the amplifier could be represented by an input resistor at temperature T. Fine in concept, but not very practical.
Applying a little maths, we do have a practical measurement method which is known as the Y factor method. It involves measuring the ratio of noise power output (Y) for two different source resistor temperatures, Tc and Th. We can say that \(NF=10 log \frac{(\frac{T_h}{290}-1)-Y(\frac{T_c}{290}-1)}{Y-1}\).
AN 57-1 contains a detailed mathematical explanation / proof of the Y factor method.
We can buy a noise source off the shelf, they come in a range of hot and cold temperatures. For example, one with specified Excess Noise Ratio (a common method of specifying them) has Th=9461K and Tc=290K. If we measured a DUT and observed that Y=3 (4.77dB) we could calculate that NF=12dB.
This method of noise figure measurement is practical and used widely. Note that the DUT always has its nominal terminations applied to the input and output, the system gain is maintained, just the input equivalent noise temperature is varied.
Some amplifiers are not intended to be impedance matched at the input (ie optimised for maximum gain), but are optimised for noise figure by controlling the source impedance seen at the active device. Notwithstanding that the input is not impedance matched, noise figure measurements are made in the same way as for a matched system as they figures are applicable to the application where for example the source might be a nominal 50Ω antenna system.
So, NF is characterised for an amplifier with its intended / nominal source and load impedances.
Nothing about the NF implies the equivalent internal noise with a short circuit SC or open circuit OC input. The behaviour of an amplifier under those conditions is internal implementation dependent (ie variable from one amplifier design to another) and since it is not related to the amplifier’s NF, it is quite wrong to make inferences based on noise measured with SC or OC input.
So this raises the question of NF measurements made with a 50Ω source on an amplifier normally used with a different source impedance, and possibly a frequency dependent source impedance. An example of this might be an active loop amplifier where the source impedance looks more like a simple inductor.
Well clearly the measurement based on a 50Ω source does not apply exactly as amplifier internal noise is often sensitive to the source impedance, but for smallish departures, the error might be smallish.
A better approach might be to measure the amplifier with its intended source impedance. In the case of the example active loop antenna, the amplifier could be connected to a dummy equivalent inductor, all housed in a shielded enclosure and the output noise power measured with a spectrum analyser to give an equivalent noise power density at the output terminals. Knowing the AntennaFactor of the combination, that output power density could be referred to the air interface. This is often done and the active antenna internal noise expressed as an equivalent field strength in 1Hz, eg 0.02µV/m in 1Hz. For example the AAA-1C loop and amplifier specifies Antenna Factor Ka 2 dB meters-1 @ 10 MHz
and MDS @ 10MHz 0.7 uV/m , Noise bandwidth =1KHz and
to mean equivalent internal noise 0.022µV/m in 1Hz @ 10MHz at the air interface. 0.022µV/m in 1Hz infers Te=6.655e6K and NF=43.608dB again, at the air interface. These figures can be used with the ambient noise figure to calculate the S/N degradation (SND).
A spectrum analyser or the like can be used to measure the total noise power density at the output of the loop amplifier with the input connected to a dummy antenna network (all of it shielded) and to calculate the equivalent noise temperature and noise figure at that point. For example, if we measured -116dBm in 1kHz bandwidth, Te=1.793e+5K and NF=27.9dB. Knowledge of the gain from air interface to that reference point is needed to compare ambient noise to the internal noise and to calculate SND, that knowledge might come from published specifications or a mix of measurements and modelling of the loaded antenna.
The mention of a spectrum analyser invites the question about the suitability of an SDR receiver. If the receiver is known to be calibrated, there is no non-linear process like noise cancellation active, and the ENB of the filter is known accurately, it may be a suitable instrument.
In both cases, the instruments are usually calculated for total input power, ie external signal and noise plus internal noise, so to find external noise (ie from the preamp) allowance must be made for the instrument NF (ie it needs to be known if the measured power is anywhere near the instrument noise floor).
Field strength / receive power converter may assist in some of the calculations.
The foregoing discussion assumes a linear receiver, and does not include the effects of intermodulation distortion IMD that can be hugely significant, especially in poor designs.
Part of the problem of IMD is that the effects depend on the individual deployment context, one user may have quite a different experience to another.
There are a huge number of published active loop antenna designs and variant, and a smaller number of commercial products. Most are without useful specifications which is understandable since most of the market are swayed more by anecdotal user experiences and theory based metrics and measurement.
Online posters are excited that it supports some versions of nanoVNA, and one thread attempts to answer the questions:
The SWR image shows that the SWR minimum is at the center phase angle as you would expect. My question is:
what are the other points that look like resonance,
and should I trim my antenna based on phase?
If so which one?
They are interesting questions which hint the ham obsession with resonance as an optimisation tartget.
Properly interpreting VNA or analyser measurements starts with truly understanding the statistic being interpreted.
In this case, the statistics being discussed are Phase and VSWR, and their relationship.
What is the Phase being discussed?
Above is an Antscope2 phase plot for an archived antenna measurement. The measurements are of a 146MHz quarter wave mobile antenna looking into about 4m of RG58C/U cable. We will come back to this.
Above is the SWR (VSWR) plot.
Let’s analyse both of these.
What is VSWR?
Well you do need to understand the meaning of VSWR, the context, and expected values for the type of antenna being measured.
Minimum VSWR occurs at about 147.625MHz in this case, and at 1.3, it is quite within expectations for this type of antenna.
What is phase?
Let’s assume the reported complex Z at the cursor in the phase plot is correct, and calculate some interesting values from it.
Above, the phase of Γ is -160.6° which reconciles with the plotted phase… so it would seem that the plot of phase is actually a plot of the phase of the complex reflection coefficient Γ.
Is that useful in the context of the questions posed above about optimising VSWR and resonance?
No.
Above is the phase plot from Antscope (1) for the same file.
Phase is reported at 147.625 as -11.6°.
Can Antscope(1) and Antscope2 be plotting the same thing on their respective phase plots?
No. Antscope(1) plots the argument of Z and Antscope2 plots the argument of Γ.
Are either of them directly useful to determine if the antenna itself is resonant?
No. We are looking into a feed line, and without de-embedding the feed line from the measurements, neither phase directly indicates resonance of the antenna itself.
In fact, for a quarter wave ground plane antenna, minimum VSWR occurs very close to resonance, and feed line losses are least at minimum VSWR so it should be the optimisation target.
]]>Warned of a potential quality issue, I measured my own AA-600.
Above, the test of the inner pin forward surface distance from the reference plane on the N jack on the AA-600. The acceptable range on this gauge for the female connector is the red area, and it is comfortably within the red range.
Above is a table of critical dimensions for ‘ordinary’ (ie not precision) N type connectors from Amphenol.
This dimension is important, as if the centre pin protrudes too much, it may damage the mating connector.
Pleased to say mine is ok, FP at 0.192″.
I used a purpose made gauge to check this, but it can be done with care with a digital caliper (or dial caliper or vernier caliper), that is what I did for decades before acquiring the dial gauge above.
]]>quite normal, this article suggests there is an obvious possible explanation and that to treat it as quite normal may be to ignore the information presented.
Above is a partial simulation of a scenario using Rigexpert’s Antscope. It starts with an actual measurement of a Diamond X-50N around 146MHz with the actual feed line de-embedded. Then a 100m lossless feed line of VF=0.66 is simulated to produce the plot that contains a ripple apparently superimposed on an expected V shaped VSWR curve.
This is the type of ripple that the expert’s opine is quite normal.
Let’s apply some thinking to this.
The ripple is periodic with frequency, the pattern repeats about every 1MHz around 146MHz, or about 0.68% of frequency, or 1/146 of frequency.
The question that should be asked is, is this periodicity related to the electrical length of the feed line? Recall that VSWR patterns repeat every half wavelength along a transmission line, so we are interested in the feed line length in half wavelengths.
The feed line is 73.8λ at 146MHz, or 147.6 half wavelengths. This means that with a frequency change of 1/147.6, we will increase the electrical length by one half wavelength.
In this case, the frequency periodicity of the plotted VSWR curve is approximately the same as the frequency periodicity of the electrical length of the line, and is a likely explanation of the VSWR curve if the reference impedance for the VSWR plot is significantly different to the characteristic impedance of the transmission line section.
Since Zref of the instrument is 50Ω we might properly suspect that the VSWR plot ripple is caused by depart of the transmission line from 50Ω and using Antscope, explore other Zrefs to minimise the ripple.
Above is a plot with Zref=45Ω, it is more like what might be expected, and it suggests that the transmission line has Zo=45Ω and that is the cause of the ripple.
If the transmission line was specified to be 50Ω, analysis of the VSWR ripple suggests the transmission line does not meet specification and warrants a more conclusive test by placing a known frequency independent load at the far end and measuring VSWR, observing ripple and finding Zref that minimises the ripple.
Of course, the VSWR ripple may be caused because Zref of the instrument is not correct, so using an instrument that is incorrectly calibrated might produce similar ripple.
Above is a simulation where the 100m transmission line Zo=50Ω and Zref=45Ω. Check that instrument Zref properly suits the transmission line before making measurement.
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