The article is ‘in-brief’ as technical stuff that will not interest most hams is published privately on a members-only page. This article is based on the information in the QST article alone (ie not on the private members only supplementary information).

The core has a modest price in North America, but shipping to other parts of the world may make it very expensive… IOW unobtainium to most parts of the world.

Above is the published InsertionLoss. The article states that they were half the value obtained in a back to back measurement, and it should be noted that is a compromised measurement, and secondly that InsertionLoss comprises two components, (dissipative) Loss and MismatchLoss.

The fact that InsertionLoss increases markedly at the lower frequencies is a hint that it has too few turns for those frequencies, and that InsertionVSWR and efficiency may suffer.

The transformer is intended to be used with a load such that the input impedance Zin is approximately 50+j0Ω, Gin=0.02S.

Analysis of a simple model of the transformer with a load such that input impedance is 50+j0Ω gives insight into likely core losses in that matched condition.

Let us calculate the magnetising admittance of the 4t primary at 1.8MHz. The core is a Laird 28B1540-000 ferrite toroid.

Above is a table of complex permeability for Laird 28 material. (I have seen online experts advise that this information is not available… but here it is.) The Laird #28 material is a medium μ NiZn ferrite, somewhat similar to Fair-rite #43 but different enough to not be equivalent.

Now lets model the 4t primary at 1.8MHz.

Gcore is the real part of Y, 0.000826S.

If Yin of the loaded transformer is 0.02S, we can calculate the core efficiency as 1-Gcore/Gin=1-0.000826/0.02=95.8%, core loss is 0.18dB. Note that this efficiency is in the matched condition, and could be higher or lower at other input impedances (η=1-Gcore/Gin). Ferrite materials have fairly wide tolerance, so measurements of actual cores may have some variation.

The balance of the measured InsertionLoss is likely to be MismatchLoss, but that information is not disclosed in the main article. MismatchLoss will include contributions from the shunt magnetising admittance (moderately low in this case) and effects of flux leakage.

- Find |Z|,R,|X| from VSWR,|Z|,R,Ro
- A new impedance calculator for RF inductors on ferrite cores
- Calculate ferrite cored inductor (from Al)
- Calculate VSWR and Return Loss from Zload (or Yload) and Zo
- Duffy, O. 2015. A method for estimating the impedance of a ferrite cored toroidal inductor at RF. https://owenduffy.net/files/EstimateZFerriteToroidInductor.pdf.
- ———. 2006. A method for estimating the impedance of a ferrite cored toroidal inductor at RF. VK1OD.net (offline).

]]>

AIM915a was recently pulled from the distribution site and replaced by a new release, AIM916.

AIM916 chokes on some calibration files created with earlier versions, so again historical scan data is rendered worthless. Note the illogical diagnostic message… typical AIM quality.

I cannot recall ever finding a new release that did not have significant defects, commonly inconsistency between displayed values. In the common theme of one step forward, two steps backwards, this version has defects that were not present in AIM910B.

This problem existed in AIM915a, it persists in AIM916.

Let’s review the internal consistency of this part of the display screen.

Most of the values given above are calculated from a single measurement value, and should be internally consistent. That measurement value is translated to different quantities, many based on the stated Zref (50Ω in this case).

Let’s assume that the values given for Rs and Xs are the basis for the other quantities, and check the consistency of those other values with Rs, Xs based on Zref=50Ω as shown.

Above are various values calculated from Rs=49.720Ω, Xs=0.090Ω and Zref=50+j0Ω.

Working down from the top of the AIM results:

- SWR (1.508) is wrong, it should be 1.01;
- Rho Mag (0.2027) is wrong, it should be 0.002949;
- s11 (-0.0028+j0.0009) is ok;
- % refl power (4.1) is wrong, it should be 0.00087;
- Return Loss (13.86dB) is wrong, it should be 50.61B; and
- Cable Loss (6.93dB) is wrong, it should be 25.30dB.

What was fixed in AIM916? Well, that seems to be a secret… but some stuff that worked was broken (the cal file issue) and some stuff that was broken remains broken.

Does not inspire a lot of confidence, does it?

Will this internal error that has been in many released versions ever get fixed before the hardware is so dated as to be worthless anyway? Some might say that time has already passed… newer competitive hardware with greater ADC resolution obsoletes the AIMuhf.

]]>This article reviews the magnetics design of the -20dB / 25W coupler.

The coupler uses a type of Sontheimer coupler (Sontheimer 1966) and these are commonly poorly designed. The first question is whether the magnetising impedance of T2 which appears in shunt with the load is sufficiently high to not give rise to poor insertion VSWR.

The instructions specify a FT50-43 with 10t for 1MHz and up. Experience says this is probably low turns and worth reviewing.

Though rated down to 1MHz, this review is at 1.8MHz, the lowest ham band above 1MHz. (It has worse InsertionVSWR at 1MHz.)

Let’s calculate the expected magnetising admittance.

Magnetising Y=0.000598-j0.00241S, let’s calculate VSWR with 50Ω in shunt, so Yt=0.020598-j0.00241S.

Ok, 10t gives rise to an InsertionVSWR of 1.13 at 1.8MHz considering the core alone, which is on the poor side.

It is worth reviewing the expected dissipation of the T2 core as this is a failing of many designs.

At 25W in a 50Ω system, V=(25*50)^0.5=35V. The power dissipated in the T2 core is at Pfwd=25W is calculated as E^2*G=35^2*0.000598=0.73W. The core is likely to get quite warm, but in free air should dissipate that heat comfortably.

The performance may be better with a core with double the ΣA/l (or double Al).

The 30dB coupler can be analysed using the same method. It is left as an interesting exercise for the reader.

- Sontheimer,C & Frederick,RE. Apr 1966. Broadband directional coupler. US Patent 3,426,298.

AIM915 was recently pulled from the distribution site and replaced by a new release, AIM915a.

I cannot recall ever finding a new release that did not have significant defects, commonly inconsistency between displayed values. In the common theme of one step forward, two steps backwards, this version has defects that were not present in AIM910B.

This problem existed in AIM915, it persists in AIM915a.

Let’s review the internal consistency of this part of the display screen.

Most of the values given above are calculated from a single measurement value, and should be internally consistent. That measurement value is translated to different quantities, many based on the stated Zref (50Ω in this case).

Let’s assume that the values given for Rs and Xs are the basis for the other quantities, and check the consistency of those other values with Rs, Xs based on Zref=50Ω as shown.

Above are various values calculated from Rs=49.719Ω, Xs=0.090Ω and Zref=50+j0Ω.

Working down from the top of the AIM results:

- SWR (1.508) is wrong, it should be 1.01;
- Rho Mag (0.2027) is wrong, it should be 0.002959;
- s11 (-0.0028+j0.0009) is ok;
- % refl power (4.1) is wrong, it should be 0.00087;
- Return Loss (13.86dB) is wrong, it should be 50.58dB; and
- Cable Loss (6.93dB) is wrong, it should be 25.29dB.

Does not inspire a lot of confidence, does it?

]]>AIM914 was recently pulled from the distribution site and replaced by a new release, AIM915.

I cannot recall ever finding a new release that did not have significant defects, commonly inconsistency between displayed values. In the common theme of one step forward, two steps backwards, this version has defects that were not present in AIM910B.

Let’s review the internal consistency of this part of the display screen.

Most of the values given above are calculated from a single measurement value, and should be internally consistent. That measurement value is translated to different quantities, many based on the stated Zref (50Ω in this case).

Let’s assume that the values given for Rs and Xs are the basis for the other quantities, and check the consistency of those other values with Rs, Xs based on Zref=50Ω as shown.

Above are various values calculated from Rs=49.719Ω, Xs=0.090Ω and Zref=50+j0Ω.

Working down from the top of the AIM results:

- SWR (1.508) is wrong, it should be 1.01;
- Rho Mag (0.2027) is wrong, it should be 0.002959;
- s11 (-0.0028+j0.0009) is ok;
- % refl power (4.1) is wrong, it should be 0.00087;
- Return Loss (13.86dB) is wrong, it should be 50.58dB; and
- Cable Loss (6.93dB) is wrong, it should be 25.29dB.

Does not inspire a lot of confidence, does it?

]]>Of interest to European designers is inclusion of three common materials used for HF applications, Ferroxcube’s 4A11, 4B2, and 4C65.

]]>An online expert questioned the analysis and later measurements, and proposed his own transformer design as evidence.

Notably, his transformer uses #61 material and a larger binocular core, a Fair-rite 2861006802 with 2t for a nominal 50Ω primary, giving loss measurements at 7MHz of 0.08dB. Note that the confidence limits of that loss measurement because of the way in which it was obtained (eg a 1% error in the 1120Ω load resistor contributes 0.043dB error to the result), but the measurements do suggest that the loss is probably very low.

Though the loss is low and Return Loss is high at 7MHz, the limits for ReturnLoss>14dB (VSWR<1.5) is 5-18MHz. With compensation, that range may be changed.

Lets apply the method laid out at PD7MAA’s BN43-202 matching transformer for an EFHW.

The best Fair-rite data I can find quickly is a chart of the impedance of a one turn winding.

Scaling from this graph, Xs is close of 35Ω at 7MHz, so lets used that to derive some basic parameters for the core.

Firstly, lets find the permeability of #61 at 7MHz.

Freq (MHz) |
µ’ |
µ” |

7.000e+0 |
1.214e+2 | 1.159e+0 |

Using that in a calculator to iteratively find the value of ΣA/l that gives Xs=35Ω at 7MHz, we obtain ΣA/l=0.0054m, this captures the magnetic path geometry of the binocular core.

Let us now use that core characteristic to calculate the magnetising admittance of the 2t primary winding.

Gcore is the real part of Y, 0.0000421S.

If Yin of the loaded transformer is 0.02S, we can calculate the core efficiency as 1-Gcore/Gin=1-0.0000421/0.02=99.8%, core loss is 0.009dB.

The total loss of this type of transformer will be dominated by the core loss.

The posted measured results, though having wide confidence limits, fall quite in line with a theoretical prediction using the method laid out at PD7MAA’s BN43-202 matching transformer for an EFHW. The measurements are evidence that the design method works.

The superior efficiency of the tested transformer is due to a better magnetic design, better than the PD7MAA design.

- PD7MAA EFHW antenna for 40-10m qrp
- Find |Z|,R,|X| from VSWR,|Z|,R,Ro
- A new impedance calculator for RF inductors on ferrite cores
- Calculate ferrite cored inductor (from Al)
- Calculate VSWR and Return Loss from Zload (or Yload) and Zo
- Duffy, O. 2015. A method for estimating the impedance of a ferrite cored toroidal inductor at RF. https://owenduffy.net/files/EstimateZFerriteToroidInductor.pdf.
- ———. 2006. A method for estimating the impedance of a ferrite cored toroidal inductor at RF. VK1OD.net (offline).

]]>

A new release, AIM914 appeared recently.

In the common theme of one step forward, two steps backwards, this version has defects that were not present in AIM910B.

Let’s review the internal consistency of this part of the display screen.

Most of the values given above are calculated from a single measurement value, and should be internally consistent. That measurement value is translated to different quantities, many based on the stated Zref (75Ω in this case).

Let’s assume that the values given for Rs and Xs are the basis for the other quantities, and check the consistency of those other values with Rs, Xs based on Zref=75Ω as shown.

Above are various values calculated from Rs=12.428Ω, Xs=17.173Ω and Zref=75+j0Ω.

Working down from the top of the AIM results:

- SWR (4.596) is wrong, it should be 6.36;
- Rho Mag (0.6426) is wrong, it should be 0.7282;
- s11 (-0.6530+j0.3251) is ok;
- % refl power (41.3) is wrong, it should be 53.0;
- Return Loss (3.84dB) is wrong, it should be 2.75dB; and
- Cable Loss (1.92dB) is wrong, it should be 1.38dB.

Does not inspire a lot of confidence, does it?

]]>Product with apparently similar specifications are sold by many ham retailers, they may or may not be sourced from Seminole.

Some sellers specify the % ICAS rating of the copper clad conductor, usually 30%, some just don’t mention it.

John carefully measured the DC resistance of his line section, and found that it reconciled well with the Copperweld datasheet for 21% CCS.

He also used a VNA to measure S11 of the line section with S/C and O/C terminations, and he gives links to the Touchstone files at the top of his page.

The O/C Touchstone file allows calculation of Zin. The O/C line exhibits resonance at 4.2MHz, at Zin=3.7Ω. His fuller set of measurements showed that Zo at 4.2MHz is very close to 400Ω. We can use those measurements to calculate Matched Line Loss (MLL).

Above, MLL is 0.50852dB/100m.

A graph was made of:

- MLL calculated above from KN5L’s measurements for 21% ICAS line;
- MLL inferred by the physical parameters for 21%, 31% and 100% cladding as described at A model of current distribution in copper clad steel conductors at RF;
- An extrapolation of N7WS’s measurements of Wireman 551 at 50MHz (where it behaves essentially as a 100% copper conductor). The extrapolation is unsafe because the conductors are unlikely to be as good as copper at 4.2MHz, and this will be an underestimate of loss; and
- TLDetails estimate of MLL for Wireman 551, a similar line though specified as 30% ICAS.

Above, the modelled current distribution in the conductor.

Above, the different values charted for comparison.

The closest value to the measurement is that found from the 21% CCS current distribution model.

The furthest from measured is TLDetails, and although it is for a 30% ICAS conductor, it gives 65% higher MLL.

The 100% copper conductor model reconciles very well with extrapolation of N7WS’s measurements.

We can reasonably draw the conclusion that the 21% CCS line has MLL a little higher than the 100% copper equivalent at 4.2MHz. Note that this is a single core conductor, stranded 21% CCS is likely to be worse.

John’s measurement raises questions for the technical audience of the ICAS specification of market products, and then their RF performance at the lower end of HF and MF.

]]>A recent article questioned the accuracy of measurement of Matched Line Loss (MLL) for a modified commercial transmission line. The published results were less than half the loss of an equivalent line in air using copper conductors and lossless dielectric, when in fact there would be good reason to expect that the line modification would probably increase loss.

How do you avoid the pitfalls of using analysers and VNAs to measure line loss?

Lets walk through a simple exercise that you can try at home with a good one port analyser (or VNA). Measuring something that is totally unknown does not provide an external reference point for judging the reasonableness of the results, so will use something that is known to a fair extent,

For this exercise, we will measure the Matched Line Loss (MLL) of a 6m length of uniform transmission line, RG58C/U cable, using an AIMUHF analyser. The AIM manual describes the method.

If you need to know the cable loss at other frequencies, enable the Return Loss display using the Setup menu and click Plot Parameters -> Return Loss and then do a regular scan of the cable over the desired frequency range with the far end of the cable open. Move the blue vertical cursor along the scan and the cable loss will be displayed on the right side of the graph for each frequency point

Note the one-way cable loss is numerically equal to one-half of the return loss. The return loss is the loss that the signal experiences in two passes, down and back along the open cable.

Our measurements will show that this is a naively simple explanation, and to take it literally as complete may lead to serious errors. Yes, it IS the equipment manual, but it is my experience that the designers of equipment, and writers of the manuals often show only a superficial knowledge of the relevant material.

Above is an extract of the datasheet for Belden 8262 RG58C/U type cable, our test cable should have similar characteristics.

This measurement is literally that described in the AIM manual.

Above, the Return Loss Plot of 6m of RG58C/U with O/C termination. (The AIM Return Loss plot is upside down, Return Loss increasing downwards, just to be quirky.)

Note that Return Loss at 1MHz is almost zero, 0.02dB and implies MLL is 0.01dB for 6m, 0.166dB/100m which is grossly different to the datasheet’s 1.3dB/100m.

How is it possible to have a similar cable with 13% of the specification MLL?

We can learn more from failures than successes, lets follow the opportunity.

Let’s examine in detail two frequencies in the sweep range, 8.2MHz (the first resonance of the section) and 3.5MHz (a typical frequency for ham use).

Looking at the screenshot above, the data at the right shows Return Loss as 8.2MHz is 0.61dB, and the calculated Cable Loss (meaning MLL) is 0.30dB.

The previous case should make us wary and test the accuracy of this result.

0.30dB/6m is equivalent to 5dB/100m, a little higher than the interpolation of the datasheet values, but believable.

Another method is to estimate loss from the input resistance of a resonant section.

From the screenshot, Rin is 1.75Ω. For the moment, lets assume that Zo is actually 50+j0Ω.

Using Calculate transmission line Matched Line Loss from Rin of o/c or s/c resonant section.

We obtain the result above, that MLL is 0.051dB/m, or 5.1dB/100m. This reconciles with the measurement, and is close to the datasheet interpolation.

On account of the fact that the datasheet, the O/C Half Return Loss and Rin of a resonant section values are very close, we conclude that both measurements are very likely to be valid.

Lets repeat that but with a scan of the same cable terminated with a S/C (though it is not mentioned in the AIM manual).

Above is the S/C sweep. Note the very different shape of the plot.

Nevertheless, at 8.2MHz, the Return Loss is almost identical, looking at the screenshot above, the data at the right shows Return Loss as 8.2MHz is 0.61dB, and the calculated Cable Loss (meaning MLL) is 0.31dB.

The previous case should make us wary and test the accuracy of this result.

0.30dB/6m is equivalent to 5dB/100m, a little higher than the interpolation of the datasheet values, but believable.

Another method is to estimate loss from the input resistance of a resonant section.

Maximum Rin around 8.2MHz is 1400Ω, so we will use that. For the moment, lets assume that Zo is actually 50+j0Ω.

Using Calculate transmission line Matched Line Loss from Rin of o/c or s/c resonant section.

We obtain the result above, that MLL is 0.052dB/m, or 5.2dB/100m. This reconciles with the measurement, and is close to the datasheet interpolation, and to the previous measurements.

We have stronger reason to conclude that all measurements are very likely to be valid.

Lets move the cursor on the O/C scan down to 3.5MHz.

Looking at the screenshot above, the data at the right shows Return Loss as 3.5MHz is 0.09dB, and the calculated Cable Loss (meaning MLL) is 0.04dB.

0.04dB/6m is equivalent to 0.67dB/100m, a lot higher than the interpolation of the datasheet values, unbelievable.

This is not a resonant section, so we cannot use the Rin method directly, but since MLL is almost entirely due to conductor loss for this type of cable at this frequency, we can interpolate from the datasheet MLL=(3.5/10)*4.59=2.71dB, or extrapolate from our own Rin based MLL at 8.2MHz MLL=(3.5/8.2)*5.1=3.3dB. Neither of these support the measurement of 0.67dB/100m… so it is probably wrong.

Lets move the cursor on the O/C scan down to 3.5MHz.

Looking at the screenshot above, the data at the right shows Return Loss as 3.5MHz is 0.59dB, and the calculated Cable Loss (meaning MLL) is 0.29dB.

0.29dB/6m is equivalent to 4.83dB/100m, a lot higher than the interpolation of the datasheet values, unbelievable.

This is not a resonant section, so we cannot use the Rin method directly, but since MLL is almost entirely due to conductor loss for this type of cable at this frequency, we can interpolate from the datasheet MLL=(3.5/10)*4.59=2.71dB, or extrapolate from our own Rin based MLL at 8.2MHz MLL=(3.5/8.2)*5.1=3.3dB. Neither of these support the measurement of 4.83dB/100m… so it is probably wrong.

The average of the O/C and S/C Return Loss is (0.09+0.59)/2=0.340, MLL=0.170/6m or 2.83dB/100m. This falls in the range between our datasheet interpolation and extrapolation from our 8.2MHz Rin based MLL.

Stated earlier was an assumption that Zo=50+j0Ω, and we often assume that Zo equals the nominal Zo for convenience… but:

- are results sensitive to Zo; and
- is the convenience safe?

For this example, I have measured complex input Z of the O/C and S/C sections at 3.5MHz. Zo can be calculated Zo=(Zoc*Zsc)^0.5.

Here is the calculation in Python

>>> zo=cmath.sqrt((0.616-61.496j)*(2.779+39.865j)) >>> zo (49.55229895635798-1.4766271906868134j)

So, Zo=49.55-j1.48Ω. (Of course this value is subject to measurement error, but it is quite within the range of expectations for this cable type at this frequency… but quantifying Zo is a larger subject in its own right.) This value of Zo is not a defect, this is a natural consequence of mainly conductor based loss at this frequency. It is this departure from nominal Zo (50+j0Ω) that gives rise to the difference between Return Loss of the O/C and S/C sections at 3.5MHz.

As explained at On Witt’s calculation of Matched Line Loss from Return Loss, taking half of the average of Return Loss for O/C and S/C sections gives a reasonably good approximation of MLL provided the error in assumed Zo is small.

If you were to make a single measurement at 3.5MHz following the method given in the AIM manual, you would obtain MLL of about 25% of the correct value, a gross error in anyone’s terms.

By comparing that value with expectation based on the datasheet, and measurements at other frequencies including using other techniques, we challenged the validity of the measurement, and ultimately the AIM manual advice,

By questioning the validity of the first measurement, we learned that:

- the Half Return Loss method is sensitive to accuracy of Zo;
- the AIM method is naively incomplete;
- Zo in the example was sufficiently different from assumed nominal Zo to cause large error in calculated MLL at 3.5MHz using the AIM method;
- estimating MLL as half the average of Return Loss for O/C and S/C sections gave a reasonably good approximation of MLL provided the error in assumed Zo was small.