For this discussion, I will use the amplifier developed at A high performance active antenna for the high frequency band, but applied to the antenna described at Ambient noise measurement using whip on vehicle – #1 – estimate Antenna Factor.

Let’s assume that the antenna + amplifier will be used with a HF receiver with Noise Figure 6dB, Teq=864.5K.

From (Martinsen 2018) Fig 3.8, the amplifier internal noise at the output terminals is -118dBm in 100kHz @ 3.5MHz. That implies that the amplifier Noise Temperature is 857.93K. The amplifier has 6.4dB voltage gain which needs to be subtracted from the AF calculated for unity gain (at the amplifier input terminals).

Amplifier characteristics:

- Voltage gain 6.4dB;
- AF=6.05-6.4=-0.35dB/m;
- input Z=1MΩ||15pF;

From (Martinsen 2018) Fig 3.8, the amplifier internal noise at the output terminals is -118dBm in 100kHz @ 3.5MHz. The equivalent noise temperature is 857.9K.

The amplifier output terminals will be used as the reference plane for the following calculations.

Total internal noise (amplifier and receiver) \(T’=857.9+864.5=1722 \text{ K}\).

Using Field strength / receive power converter for measurements made in a 1kHz wide ENB,

we obtain:

The Ambient Noise Figure Fa is 57.5dB, similar to that predicted for a Residential precinct in ITU P.372-14.

It may be tempting to assume a simple linear relationship between the measure power and ambient noise figure.

In most cases, the measured power includes internal noise of the receive system (both active antenna and the receiver), and when external noise dominates, the relationship is close to linear.

Above is a plot of Fa vs measured (total) power for this (exact) scenario. In this case, Fa=Pt+174 is a good estimator for Pt>-145dBm or Fa>29dB. You would not expect to measure such low Fa in this scenario (frequency and equipment), but that may not be true of other scenarios… you need to check.

Field strength / receive power converter properly accounts for internal noise.

- Martinsen, W. Aug 2018. A high performance active antenna for the high frequency band. Cyber and Electronic Warfare Division

Defence Science and Technology Group DST-Group-TR-3522.

Ambient noise is commonly dominated by man made noise, and it often arrives equally from all directions. For measurement of such noise, the captured power depends on average antenna gain, and so the calculations below focus on gain averaged over the hemisphere.

Antenna Factor is often very convenient for field strength measurement as it relates the external E field strength to the receiver terminal voltage given a certain antenna (system). In fact, given a short vertical terminated by a high impedance amplifier, Antenna Factor is often fairly independent of frequency over several octaves of frequency.

Whilst it is easy to come up with Rules of Thumb or simple approximations for a short monopole over perfect earth conductor (PEC) either matched for maximum power transfer, or essentially unloaded, the case of a short vertical on the roof of a motor vehicle suspended above natural ground is not so easy.

Some underlying assumptions:

- a vertically polarised antenna will be most sensitive to ground wave noise on the lower HF bands (as horizontally polarised ground waves are more quickly attenuated with distance);
- the system uses an active antenna, the amplifier having a high input impedance and unity voltage gain; and
- reciprocity can be used to infer certain receive performance from transmit performance.

So, the approach is to:

- calculate the power captured by a lossless isotropic antenna, and the Thevenin source voltage for a given E field (1V/m);
- use an NEC model to find the (matched) average gain of the antenna system (so accounting for ground losses etc), source impedance, and radiation resistance (which is used as the Thevenin source impedance); and
- calculate the loaded voltage at the amplifier input terminals, and Antenna Factor at the amplifier input terminals.

An NEC-5.0 model was created for a 1.215m (4′) vertical mounted on the roof of a motor vehicle at 3.5MHz. The model was derived from a sample model supplied with NEC-5, the vertical was lengthened and moved to mid roof, and the unused antennas deleted. The model was changed to introduced real ground (σ=0.005, εr=13).

Above is a 3D pattern plot, the pattern is an almost omnidirectional donut.

Above is an elevation profile.

Above is the azmuth plot at θ=-70° (elevation 30°).

Some key values are extracted from the NEC output report.

The calculated equivalent series source capacitance Cs is given for interest sake, it is not used directly here, but is often estimated for so-called E-field probe antennas.

Thevenin source voltage of a lossless isotropic antenna is found by calculating the available power to be captured by a matched antenna given the excitation scenario (E=1V/m). The available power is given by the product of the effective aperture Ae and the power flux density S (for E=1V/m). Ae is calculated from average gain (unity for a lossless isotropic antenna) and frequency.

Having calculated available power, we can calculate the voltage in a matched load, and the Thevenin source voltage Vth is twice that.

The Thevenin source impedance is radiation resistance Rr calculated earlier from the NEC output.

We are interested in the Antenna Factor when the antenna is loaded by the high impedance amplifier. The amplifier is approximated as some resistance Rp in shunt with some capacitance Cp.

Antenna Factor is given by \(AF=20 log \frac{E}{V_{in}} \text{ dB}\) where E is the electric field strength and Vin is the amplifier input terminal voltage.

AF is quite sensitive to Cp, efforts need to be made in circuit configuration, device selection, and circuit layout to minimise Cp and achieve optimal AF.

The above discussion is based on a unity gain amplifier, if otherwise, the gain in dB should be subtracted from the calculated AF to get AF wrt the amplifier output terminals (ie, the input terminals to the following receiver).

]]>Above is an archived extract of a spreadsheet that was very popular in the ham community, both with antenna designers and sellers and end users (buyers / constructors). It shows a column entitled G/T which is actually the hammy calculation. The meaning possibly derives from (Bertelsmeier 1987), he used G/Ta.

Ta is commonly interpreted by hams to be Temperature – antenna. It is true that antennas have an intrinsic equivalent noise temperature, it relates to their loss and physical temperature and is typically a very small number. But as Bertelsmeier uses it, it is Temperature – ambient (or external), and that is how it is used in this article.

Let’s calculate the G/Ta statistic for the three scenarios in Do I ‘need’ a masthead preamp to work satellites on 2m? – space noise scenario.

Above is a calculation of the base scenario, G/T=-29.74dB/K.

Also shown in this screenshot is G/Ta=-23.98dB/K.

Above is a calculation of the masthead amplifier scenario, G/T=-25.21dB/K.

Also shown in this screenshot is G/Ta=-23.98dB/K.

Above is a calculation of the LNA at the receiver scenario, G/T=-25.754dB/K.

Also shown in this screenshot is G/Ta=-23.98dB/K.

Scenario | G/T (dB/K) | G/Ta (dB/K) |

Base | -29.74 | -23.98 |

With masthead LNA Gain=20dB NF=1dB | -25.21 | -23.98 |

With local LNA Gain=20dB NF=1dB | -25.75 | -23.98 |

Note that G/Ta is the same for all three configurations, it does not contain the important information that differentiates the performance of the three configurations.

Importantly, you cannot derive G/T from G/Ta without knowing either G or Ta (and some other important stuff), the G/Ta figure by itself cannot be ‘unwound’… so if you select an antenna ranked on a G/Ta value (even if mislabeled), the ranking of ‘real’ G/T may be different depending on many factors specific to your own scenario, ie the one with the better G/Ta might have the poorer G/T.

- Bertelsmeier, R. 1987. Equivalent noise temperatures of 4-Yagi-arrays for 432MHz. DUBUS..
- Duffy, O. 2006. Effective use of a Low Noise Amplifier on VHF/UHF. VK1OD.net.
- ———. 2007. Measuring system G/T ratio using Sun noise. VK1OD.net.
- ———. 2009. Quiet sun radio flux interpolations. https://owenduffy.net/calc/qsrf/index.htm.
- ITU-R. 2000. Recommendation ITU-R S.733-2 (2000) Determination of the G/T ratio for earth stations operating in the fixed-satellite service .

Base scenario is a low end satellite ground station:

- 144MHz;
- terrestrial noise (satellite with omni antenna);
- IC-9700, assume NF=4.8dB;
- omni antenna;
- 10m of LMR-400.

A metric that may be used to express the performance of an entire receive system is the ratio of antenna gain to total equivalent noise temperature, usually expressed in deciBels as dB/K. G/T is widely used in design and specification of satellite communications systems.

G/T=AntennaGain/TotalNoiseTemperature 1/K

Example: if AntennaGain=50 and TotalNoiseTemperature=120K, then \(G/T=\frac{50}{120}=0.416 \text{ } 1/K\) or -3.8 dB/K**.**

The utility of G/T is that receive S/N changes dB for dB with G/T, in fact you can calculate S/N knowing G/T, wavelength, bandwidth and the field strength of the signal (Duffy 2007).

\(Signal/Noise=S \frac{\lambda^2}{4 \pi} \frac{G}{T} \frac1{k_b B}\) where:

S is power flux density;

λ is wavelength;

k_{b} is Boltzmann’s constant; and

B is receiver equivalent noise bandwidth

Usage in this article is consistent with the industry standard meaning of G/T given at (ITU-R. 2000) (as opposed to the meaning used by some Hams who have appropriated the term for their own purpose).

Note this is not the bodgy G/T figure used widely in ham circles.

Ambient noise temperature Ta is an important factor in calculation of G/T. Ta depends on frequency, the environment, the antenna’s ability to reduce off boresight noise, and the on-boresight noise. For the purposes of this discussion let’s assume total ambient noise for the given omni satellite scenario at 144MHz is 1500K.

Above is a calculation of the base scenario, G/T=-33.41dB/K.

Above is a calculation of the masthead amplifier scenario, G/T=-31.99dB/K.

Scenario | G/T (dB/K) |

Base | -33.41 |

With masthead LNA Gain=20dB NF=1dB | -31.99 |

The first finding is that adding a masthead LNA with 20dB gain and 1dB NF makes only a small difference to G/T and hence S/N, just 1.4dB in this case.

The foregoing analysis assumed a linear receive system, no intermodulation distortion. Now let’s talk about the real world.

Some LNAs are sold without specifications, those that have meaningful NF and Gain specifications are usually based on laboratory measurements with no interfering signals.

When attached to an antenna, the out of band signals will give rise to noise due to intermodulation distortion, so the NF in-situ might be poorer than specification NF. Indeed, the IMD noise can be so great as to deliver worse G/T with the LNA.

One way of reducing IMD noise is to limit the amplitude of interfering signals arriving at the LNA active device, and front end filtering is one possible solution.

Be aware that lots of hammy Sammy LNA designs have very little front end selectivity, relying upon the narrow band response of a high gain antenna. When these are used with low gain tuned antennas, or worse, broadband antennas like Discones, the IMD noise can be huge.

On the other hand, there are LNAs available with a very narrow front end filter… but they cost a lot more.

The benefit / necessity of front end filtering depends on your own IMD scenario.

- Duffy, O. 2006. Effective use of a Low Noise Amplifier on VHF/UHF. VK1OD.net.
- ———. 2007. Measuring system G/T ratio using Sun noise. VK1OD.net.
- ———. 2009. Quiet sun radio flux interpolations. https://owenduffy.net/calc/qsrf/index.htm.
- ITU-R. 2000. Recommendation ITU-R S.733-2 (2000) Determination of the G/T ratio for earth stations operating in the fixed-satellite service .

Base scenario is a low end satellite ground station:

- 144MHz;
- satellite;
- IC-9700, assume NF=4.8dB;
- high gain (narrow beamwidth antenna);
- 10m of LMR-400.

A metric that may be used to express the performance of an entire receive system is the ratio of antenna gain to total equivalent noise temperature, usually expressed in deciBels as dB/K. G/T is widely used in design and specification of satellite communications systems.

G/T=AntennaGain/TotalNoiseTemperature 1/K

Example: if AntennaGain=50 and TotalNoiseTemperature=120K, then \(G/T=\frac{50}{120}=0.416 \text{ } 1/K\) or -3.8 dB/K**.**

The utility of G/T is that receive S/N changes dB for dB with G/T, in fact you can calculate S/N knowing G/T, wavelength, bandwidth and the field strength of the signal (Duffy 2007).

\(Signal/Noise=S \frac{\lambda^2}{4 \pi} \frac{G}{T} \frac1{k_b B}\) where:

S is power flux density;

λ is wavelength;

k_{b} is Boltzmann’s constant; and

B is receiver equivalent noise bandwidth

Usage in this article is consistent with the industry standard meaning of G/T given at (ITU-R. 2000) (as opposed to the meaning used by some Hams who have appropriated the term for their own purpose).

Note this is not the bodgy G/T figure used widely in ham circles.

Ambient noise temperature Ta is an important factor in calculation of G/T. Ta depends on frequency, the environment, the antenna’s ability to reduce off boresight noise, and the on-boresight noise. For the purposes of this discussion let’s assume total ambient noise for the given satellite scenario at 144MHz is 250K.

Above is a calculation of the base scenario, G/T=-29.74dB/K.

Above is a calculation of the masthead amplifier scenario, G/T=-25.21dB/K.

Above is a calculation of the LNA at the receiver scenario, G/T=-25.754dB/K.

Scenario | G/T (dB/K) |

Base | -29.74 |

With masthead LNA Gain=20dB NF=1dB | -25.21 |

With local LNA Gain=20dB NF=1dB | -25.75 |

The first finding is that adding a masthead LNA with 20dB gain and 1dB NF makes a small difference to G/T and hence S/N, 4.5dB in this case.

Note that there is only a small degradation in moving the LNA from masthead to local to the transceiver. There are additional reliability / maintenance issues with masthead located amplifiers… particularly if high performance narrow band front end filtering is used. It is much more practical to house a coaxial resonator (‘can’ in repeater parlance) in the shack that at the masthead.

The foregoing analysis assumed a linear receive system, no intermodulation distortion. Now let’s talk about the real world.

Some LNAs are sold without specifications, those that have meaningful NF and Gain specifications are usually based on laboratory measurements with no interfering signals.

When attached to an antenna, the out of band signals will give rise to noise due to intermodulation distortion, so the NF in-situ might be poorer than specification NF. Indeed, the IMD noise can be so great as to deliver worse G/T with the LNA.

One way of reducing IMD noise is to limit the amplitude of interfering signals arriving at the LNA active device, and front end filtering is one possible solution.

Be aware that lots of hammy Sammy LNA designs have very little front end selectivity, relying upon the narrow band response of a high gain antenna. When these are used with low gain tuned antennas, or worse, broadband antennas like Discones, the IMD noise can be huge.

On the other hand, there are LNAs available with a very narrow front end filter… but they cost a lot more.

The benefit / necessity of front end filtering depends on your own IMD scenario.

For satellite work, a low gain antenna will tend to have higher Ta by virtue of side lobe contribution, and so the improvement seen above might be diminished a little.

Terrestrial ambient noise is much higher, and the improvement would be considerably less. Likewise for an omni satellite antenna. In both cases, the improvement in G/T might be less than 1dB with the same masthead LNA… download the spreadsheet and explore.

As mentioned Ta is frequency dependent, so the case for 432MHz might be quite different than the above case. In particular, the choice of masthead mounting becomes clearer on higher frequencies.

- Duffy, O. 2006. Effective use of a Low Noise Amplifier on VHF/UHF. VK1OD.net.
- ———. 2007. Measuring system G/T ratio using Sun noise. VK1OD.net.
- ———. 2009. Quiet sun radio flux interpolations. https://owenduffy.net/calc/qsrf/index.htm.
- ITU-R. 2000. Recommendation ITU-R S.733-2 (2000) Determination of the G/T ratio for earth stations operating in the fixed-satellite service .

In fact what is commonly measured is Signal + Noise to Noise ratio, and of course this ratio is one of powers. For this reason, specifications often give (S+N)/N.

This article discusses those metrics in the context of ‘conventional’ receivers and introduces the key role of assumed bandwidth through the concept of Equivalent Noise Bandwidth..

Let’s consider the raw S/N ratio of an ideal AM detector and ideal SSB detector.

Above is a diagram of the various vector components of an AM signal with random noise, shown at the ‘instant’ of a modulation ‘valley’. The black vector represents the carrier (1V), the two blue vectors are counter rotating vectors of each of the sideband components, in this case with modulation depth 30%, and the red vector is 0.095V of random noise rotating on the end of the carrier + sideband components.

In an ideal detector, the recovered modulation is proportional to the sum of the sideband vectors, amplitude here is 0.3V, and the noise is proportional to the noise vector 0.095V.

We can calculate the S/N ratio as \(\frac{S}{N}=20 log \frac{0.3}{0.095}=10 \;dB\). You can see now why the noise voltage of 0.095V was chosen.

We can calculate the (S+N)/N ratio as \(\frac{S+N}{N}=10 log \frac{0.3^2+0.095^2}{0.095^2}=10.4 \;dB\).

AM is commonly demodulated in an envelope detector, and their departure from ideal linearity is significant.

Above is a diagram of the same ‘signal’ and the same noise as presented to an SSB detector.

The black vector represents the carrier (1V) and the red vector is 0.095V of random noise rotating on the end of the signal component.

In an ideal detector, the recovered modulation is proportional to the signal vectors, 1.0V, and the noise is proportional to the noise vector 0.095V.

We can calculate the S/N ratio as \(\frac{S}{N}=20 log \frac{1}{0.095}=20.45 \;dB\).

We can calculate the (S+N)/N ratio as \(\frac{S+N}{N}=10 log \frac{1^2+0.095^2}{0.095^2}=20.48 \;dB\).

SSB detectors are not perfect either, but circuits that are principally mixers are typically closer to ideal than AM envelope detectors on AM. Envelope detectors used with a BFO for SSB depend on incidental mixing and are quite unpredictable.

On the basis of the above, we would expect the sensitivity figure for SSB at 10dB (S+N)/N to be 10.5dB lower (or about one third) that for the AM detector.

Yet specifications for real receivers tend to give differences more in the range of 15-20dB. Why?

Commonly, the IF bandwidth for SSB is considerably less than for AM. For communications quality we might expect 3dB less, for AM broadcast quality vs communications quality SSB, we might expect more like 6dB.

In a receiver where the noise power is dominated by the front end, the amount of noise power presented to an SSB detector from a 2.5kHz IF filter is substantially less than presented to an AM detector from a 10kHz IF filter, 6dB in this case, leading to an expectation that SSB sensitivity will be 16.5dB better than AM.

Detector non-linearity might cause a different difference.

If we know the sensitivity specification and Equivalent Noise Bandwidth we can calculate the receiver Noise Figure.

The problem is that receiver specifications tend to not give the Equivalent Noise Bandwidth, rather they may give the bandwidth between nominated points like -6dB, -60dB etc and that does not imply Equivalent Noise Bandwidth.

The difference in stated sensitivity for conventional communications receivers between AM and SSB modes is due to two main contributions:

- the fact that it is the amplitude of the AM sidebands that determine the S part of S/N rather than that of the carrier which is the stated sensitivity figure; and
- the Equivalent Noise Bandwidth applied to each mode by the receiver.

Let’s work an example using Simsmith to do some of the calculations.

Scenario:

- 20m ground mounted vertical base fed against a 2.4m driven earth electrode @ 0.5MHz;
- 10m RG58A/U coax; and
- Receiver with 500+j0Ω ohms input impedance and Noise Figure 20dB.

An NEC-4.2 model of the antenna gives a feed point impedance of 146-j4714Ω and radiation efficiency of 0.043%, so radiation resistance \(Rr=146 \cdot 0.00043=0.0063\).

Above, the NEC antenna model summary.

Above is a Simsmith model of the system scenario.

R1 and G model the antenna, G uses Rr for Zo, and R1 contains the balance of the feed point impedance.

With the useZo source type, the source would deliver 1W or 0dBW to a conjugate matched load.

The next important figure is the power into the 500Ω load L. it is -58.3dBW. Simsmith has calculated the solution to the antenna loss elements, mismatches and coax loss under standing waves. Effectively, the average gain of the antenna system (everything to the right of L) is -58.3dB. Such an antenna is likely to have a Directivity of around 4dB, in fact the NEC model calculates 4.8dB. So the maximum gain is -58+4.8=-53.2dB.

The burning question is whether it is sufficiently good to hear most signals. Well, a better question is how much does it degrade off-air signal to noise ratio (S/N). All receivers degrade S/N, but how much degradation occurs in this scenario.

We need to think about the ambient noise. Lets use ITU-R P.372 for guidance on the expected median noise in a rural precint.

Above, ambient noise figure @ 0.5MHz is 75.54dB.

Now lets calculate the Signal to Noise Degradation (SND).

At 4.58 dB it is not wonderful, the weakest signals (ie those with low S/N) we be degraded significantly, stronger signals (those with high S/N) will be degraded by the SAME amount, but for instance reducing S/N from 20 to 15dB is not so significant.

Applying this to your own scenario

The information fed into the calculations included:

- Rr;
- feed point impedance;
- transmission line details;
- Rx input impedance and NF; and
- Ambient noise expectation.

To calculate your own scenario, you need to find these quantities with some accuracy.

Tools:

]]>- 1m active monopole;
- RTL-2832U v3 SDR dongle;
- RPi 3B+ running Spyserver; and
- SdrSharp client.

Above is the active whip antenna. Not optimal mounting, but as you can see from the clamps, a temporary mount but one that does not confuse results with feed line common mode contribution.

Above is the remote electronics, the RPi 3B+ and RDL SDR dongle, and underneath the power supply for the active monopole.

The initial trial was on 7MHz, and was a total failure due to extreme level of RFI from the RPi itself. The two RPi power supplies tested were noise tested on a dummy load and were OK, the noise comes from the digital signals on the RPi board. A Kenwood R-5000 receiver was connected to the active antenna, and noise floor was relatively low until the RPi was plugged in.

Above is the SdrSharp screen, extreme noise level and no signals could be heard (though a local transmitter on low power verified that the receiver was working).

The emissions from the RPi, were so high that it is really unsuitable for this purpose, it would be very difficult to reduce emissions by the needed more than 40dB.

]]>A first observation of listening to a SSB telephony signal is an excessive low frequency rumble from the speaker indicative of a baseband response to quite low frequencies, much lower than needed or desirable for SSB telephony.

The most common application of such a filter is reception of A1 Morse code.

Above is a screenshot of the filter settings.

Above is a plot of the response of the filter. It is hardly an idealised rectangular filter response. Though the response might be well suited to Morse code reception, it is an issue when measurements make assumptions about the ENB. The response is not well suited to narrowband data such as RTTY etc.

A summary of the filter response follows.

Locut 0Hz.

sox: bin_width_hz=10.000Hz

Filter -6dB response: 460-770Hz=310Hz.

ENB=224Hz with respect to gain at 610Hz (passband centre frequency).

ENB=222Hz with respect to gain at 590Hz (max gain frequency).

ENB=222Hz with respect to gain at 600Hz.

If we take the gain reference frequency to be 600Hz, there is 3.5dB less noise admitted by this filter than an idealised rectangular filter. Measurements such as the ARRL MDS that might assume 500Hz bandwidth will have 3.5dB error.

A 1000Hz filter might be well suited to narrow band data reception, many of the so-called ham digital modes.

Above is a screenshot of the filter settings.

Above is a plot of the response of the filter. It is fairly close to an idealised rectangular filter response.

There appears to be no means to offset the filter at baseband frequency.

A summary of the filter response follows.

Locut 0Hz.

sox: bin_width_hz=10.000Hz

Filter -6dB response: 110-950Hz=840Hz.

ENB=823Hz with respect to gain at 530Hz (passband centre frequency).

ENB=716Hz with respect to gain at 200Hz (max gain frequency).

ENB=800Hz with respect to gain at 500Hz.

If we take the gain reference frequency to be 500Hz, there is 0.97dB less noise admitted by this filter than an idealised rectangular filter.

Above is a screenshot of the filter settings.

Above is a plot of the response of the filter. It is fairly close to an idealised rectangular filter response.

There appears to be no means to offset the filter at baseband frequency.

A summary of the filter response follows.

Locut 0Hz.

sox: bin_width_hz=10.000Hz

Filter -6dB response: 110-2350Hz=2240Hz.

ENB=2353Hz with respect to gain at 1230Hz (passband centre frequency).

ENB=1829Hz with respect to gain at 210Hz (max gain frequency).

ENB=2255Hz with respect to gain at 1000Hz.

If we take the gain reference frequency to be 1000Hz, there is 0.27dB less noise admitted by this filter than an idealised rectangular filter.

SDR# does not appear to have a convenient facility to shift or offset the baseband response.

Above is the baseband response in 2400Hz USB mode as show in the SDR# window. Note that the response rolls off below 100Hz, whereas good conventional SSB Telephony receivers would have a 6dB response from say 250-2750Hz for a ENB of 2400Hz. The lower -6dB point for this response is 110Hz.

This leads to substantial low frequency component that is not a priority for SSB telephony, and in the case where the transmitter is band limited to 300-2700Hz, the filter admits unnecessary noise and the low end and cuts of a little of the high end. It is a hammy sammy approach where recognised speech characteristics, conventions and compatibility between transmitter and receiver are jettisoned.

The basic 1000Hz USB filter provides a response close to ideal, centred around 530Hz, and its ENB is 800Hz (-0.07dB on 1000Hz).

There appears no facility in SDR# to save a number of filter settings for later recall, so the process of configuring SDR# for measurement is a bit tedious.

My attention has been draw to the facility to drag the upper and lower limits of the IF passband, thanks Martin.

Above is an example where a 500Hz passband is centred on 1500Hz at baseband.

As soon as another mode is selected, the setting is lost and there appears no facility to save a set of settings for later recall. Note the inconsistency between the two displayed bandwidth figures.

Yes, it works but it is not convenient and not practical for save / recall of a standardised set of measurement or reception conditions.

]]>(Ikin 2016) proposes a different method of measuring noise figure NF.

Therefore, the LNA noise figure can be derived by measuring the noise with the LNA input terminated with a resistor equal to its input impedance. Then with the measurement repeated with the resistor removed, so that the LNA input is terminated by its own Dynamic Impedance. The difference in the noise ref. the above measurements will give a figure in dB which is equal to the noise reduction of the LNA verses thermal noise at 290K. Converting the dB difference into an attenuation power ratio then multiplying this by 290K gives the LNA Noise Temperature. Then using the Noise Temperature to dB conversion table yields the LNA Noise Figure. See Table 1.

The explanation is not very clear to me, and there is no mathematical proof of the technique offered… so a bit unsatisfying… but it is oft cited in ham online discussions.

I have taken the liberty to extend Ikin’s Table 1 to include some more values of column 1 for comparison with a more conventional Y factor test of a receiver’s noise figure.

Above is the extended table. The formulas in all cells of a column are the same, the highlighted row is for later reference.

A test setup was arranged to measure the noise output power of an IC-7300 receiver which has a sensitivity specification that hints should have a NF≅5.4dB. The relative noise output power for four conditions was recorded in the table below.

Ikin’s method calls for calculating the third minus second rows, -0.17dB, and looking it up in his table. In my extended table LnaNoiseDifference=-0.17dB corresponds to NF=3.10dB.

We can find the NF using the conventional Y factor method from the values in the third and fourth rows.

The result is NF=5.14dB (quite close to the expected value based on sensistivity specification).

Ikin’s so called dynamic impedance method gave quite a different result in this case, 3.10 vs 5.14dB, quite a large discrepancy.

The chart above shows the relative level of the four measurements. The value of the last two is that they can be used to determine the NF using the well established theory explained at AN 57-1.

The values in the first columns are dependent on the internal implementation of the amplifier, and cannot reliable infer NF.

- Hewlett Packard. Jul 1983. Fundamentals of RF and microwave noise figure measurement. AN 57-1
- Ikin, A. 2016. Measuring noise figure using the dynamic impedance method.