This article details configuration of RTKLIB rtknavi to work on a Win10 workstation using stunnel to connect to a NTRIP server that is only available using TLS.

The version of rtklib used is v2.4.3b33, and stunnel v5.60.

The GPS receiver used here is a U-blox LEA-6T.

GPS traffic may be falsely detected by Windows as mouse traffic, and cause havoc

If you are not using a serial mouse (and most of us do not these days) t is advisable to disable serial mouse detection at startup. You can do that with the following command in a administrator authorised Powershell.

Set-ItemProperty -path "HKLM:\SYSTEM\CurrentControlSet\Services\sermouse" -name "start" -Value 4

Reboot for it to take effect.

Above is a screenshot of the serial port options. Use an appropriate COM port for your configuration.

!UBX CFG-RATE 200 1 1 !UBX CFG-MSG 2 16 0 1 0 1 0 0 !UBX CFG-MSG 2 17 0 1 0 1 0 0 !UBX CFG-MSG 240 0 0 1 0 1 # NMEA GGA !UBX CFG-MSG 240 1 0 0 0 0 # NMEA GLL !UBX CFG-MSG 240 2 0 0 0 0 # NMEA GSA !UBX CFG-MSG 240 3 0 0 0 0 # NMEA GSV !UBX CFG-MSG 240 4 0 1 0 1 # NMEA RMC !UBX CFG-MSG 240 5 0 0 0 0 # NMEA VTG !UBX CFG-MSG 240 8 0 0 0 0 # NMEA ZDA

The above commands set the message rate, enable the raw messages needed and disable some of the default NMEA messages.

!UBX CFG-RST 0

The above command resets the GPS to avoid the binary messages continuing after shutdown of the RTKLIB connection.

The combination localhost:2102 directs the packets to stunnel running on the workstation, which in turns forwards the packets in an SSL tunnel. Use your user-id and password.

Stunnel needs the following section added to its configuration file and activated. This redirects connections to localhost:2102 to ntrip.host:443 in this case, but use the applicable destination host and port

[ntrip] client=yes accept=localhost:2102 connect=ntrip.host:443 verifyChain = yes CAfile = ca-certs.pem

Stunell needs to be directed to reload the configuration when updated.

You must start stunnel before attempting to connect with rtklib, otherwise it will fail to connect to localhost:2102. If you see that message, check that you have stunnel configured correctly and running.

This solution works with Geoscience Australia’s TLS NTRIP caster.

The stunnel solution might well work with other NTRIP client apps.

]]>The example for explanation is a common and inexpensive 5943003801 (FT240-43) ferrite core.

It helps to understand what we expect to measure.

See A method for estimating the impedance of a ferrite cored toroidal inductor at RF for an explanation.

Note that the model used is not suitable for cores of material and dimensions such that they exhibit dimensional resonance at the frequencies of interest.

Be aware that the tolerances of ferrite cores are quite wide, and characteristics are temperature sensitive, so we must not expect precision results.

Above is a plot of the uncalibrated model of the expected inductor characteristic, it shows the type of response that is to be measured. The inductor is 11t wound on a Fair-rite 5943003801 (FT240-43) core in Reisert cross over style using 0.5mm insulated copper wire.

Above is the expected impedance of the same model, but looking through 100mm of lossless 200Ω line to demonstrate that connecting wires will substantially alter the measured impedance.

Make a measurement, then reduce the connecting wires to half length, if you see a significant change in the measurement then:

- the original connections were too long; and
- the reduced connection MAY still be too long.

The prototype inductor (DUT) was wound and connected to a calibrated nanoVNA with very short connecting wires. Measurement used s11 reflection technique.

Above, one terminal is tied to the SMA threads with a zip tie, and the other wire poked into the female centre pin with a little sideways tension to ensure good contact.

Above is a smoothed capture using nanoVNA-App. Note that the self resonant frequency (SRF) in this jig is 13MHz. We will use that to calibrate the predictive model in Simsmith.

This is the latest version of nanoVNA-App (v1.1.208), and it throws memory protection exceptions.

The measurements are an average of 16 sweeps (done within nanoVNA-App), there is a little measurement noise, but the results are quite usable.

The next step is to calibrate the predictive model based on measurement. The simplest calibration is to adjust the equivalent shunt capacitance to reconcile the predicted SRF with measured. In this case, the adjusted Cse is 1.48pF.

Above is a plot of the calibrated model (solid lines) and measured (circles) using a nanoVNA.

It reconciles reasonably well given the tolerances of ferrite cores.

Above is the same calibrated model with the activation of a trace to show expected core loss at 40mA inductor (common mode) current. The core is unlikely to withstand more than about 40mA continuous current in the 14MHz band, a little more in the adjacent ones, depending on enclosure and ambient conditions etc.

The model can be adjusted to explore other configurations.

The 2631803802 (FT240-31) has the same geometry as the 5943003801 (FT240-43) above but a different material, so the same aol factor is used in the Simsmith model but a different material file is called up. Additionally the number of turns is changed to 13.

Above is a plot of the uncalibrated model of the expected inductor characteristic, it shows the type of response that is to be measured. The inductor is 13t wound on a Fair-rite 2631803802 (FT240-31) core in Reisert cross over style using 0.5mm insulated copper wire.

The next step would be to measure a real inductor, and adjust the model Cse calibrate the SRF.

… an exercise for the reader.

- file: ferrite material complex permeability CSV file;
- aol: core geometry ΣA/l (Magnetic core coefficients converter);
- n: number of turns;
- cores: number of cores in stack;
- cse: equivalent shunt capacitance to account for SRF.

This article has described how to measure the choke as a component with very short leads. Packaging a choke into an enclosure introduces connecting wires etc, and measuring the packaged choke may present a greater challenge, but understanding the basic component choke is an important step in the design process.

Likewise for coming up with a realistic power rating given materials used, the environment, and intended use (duty cycle etc).

At the end, we have measurement of a single prototype, and a calibrated / validated predictive model. The latter is the better design tool, it is the objective of the process. The predictive model is a better estimator of chokes built on commercial supply of cores, you would not base a production design on a single prototype alone.

Sample Simsmith model for download: CMCSSModel.7z . (Compressed with 7zip.)

]]>So, with that background I searched eBay for replacement batteries for a Makita drill / driver. The current pair of batteries are Chinese source, purchased about 5 years ago, label rating is 2.0Ah, and measured capacity now is 1.3-1.4Ah… which is not too bad, they were 1.7-1.8Ah when new (yes, below spec).

There were very many sellers selling a lot of two packs rated at 3.6Ah for around $34 including delivery. That is very cheap, too good to believe? Let’s rely on eBay Buyer Protection to put them to the test.

The two batteries arrived quickly and were put through charge / discharge cycles to condition them.

Above are the C/5 discharge curves for both batteries on the third cycle, capacity is 1.3Ah, just 36% of the label rating.

The pack is probably built on Sub-C cells, and the best price I can quickly find for 10 Panasonic 3.05Ah cells is over $100, so that does not look too good.

A reputable brand of after market replacement battery rated for 3.0Ah is priced at just under $100.

In terms of bang for buck, these are both around $33/Ah capacity, and possibly longer life than the purchased batteries

The cost of quality cells raises the question of whether the purchased batteries are good value if they were honestly rated at 1.2Ah.

The answer is yes (unless you need the longer endurance in a single battery)… but they were even better value.

I raised an eBay dispute since the goods were not as described, and offered to accept 60% discount as compensation for the fact they had only 60% of the paid for capacity (3.0Ah), costing around $5/Ah.

]]>The antenna comprises a square loop of 3m sides of 2mm bare copper wire, buried 20mm in the soil.

Above is the site marked out for earthworks, but excavation of a narrow slot 25mm deep. On the far side of the loop is an already installed plastic irrigation valve box for the transformer.

Above, the excavation implement… an attachment for the 62cc weed wacker which is designed for cutting neat vertical edges in the grass along paths etc. It was not very expensive, so seemed worth a trial. It worked very well.

Above, wire laid and slot backfilled. This will be watered and rolled with the mower over coming weeks to grow grass roots and settle the soil.

Above, ABS tent pegs were used to secure the corners of the wire loop.

In a couple of weeks, the feed point impedance will be measured and compared to models.

After one week of settling, including lots of rain, a quick preview with a hand held receiver is promising.

A work in progress…

]]>This article models the transformer on a nominal load, being \(Z_l=n^ 2 50 \;Ω\). Keep in mind that common applications of a 50Ω:200Ω transformer are not to 200Ω transformer loads, often antennas where the feed point impedance might vary quite widely, and performance of the transformer is quite sensitive to load impedance. The transformer is discussed here in a 50Ω:200Ω context.

Above is the prototype transformer using a 2843009902 (BN43-7051) binocular #43 ferrite core, the output terminals are shorted here, and total leakage inductance measured from one twisted connection to the other.

The prototype transformer is a 3:6 turns autotransformer with the two windings twisted bifilar.

Above is the equivalent circuit used to model the transformer. The transformer is replaced with an ideal 1:n transformer, and all secondary side values are referred to the primary side.

- Secondary side leakage inductance Lls is divided by n^2 to obtain the value primary referred leakage inductance in the circuit diagram.
- Cse is an equivalent shunt capacitance to partially model self resonance effects.
- Bm is the magnetising susceptance (calculated from other parameters).
- Gm is the magnetising conductance (calculated from other parameters).
- Llp is the primary side leakage inductance.
- Ccomp is a compensation capacitance.

A Simsmith model was built to implement the transformer model above.

- Complex core permeability is captured from a permeability data file.
- np is the number of turns on the primary.
- ratio is the turns ratio.
- cores is the number of cores in a stack.
- cse is Cse per the circuit diagram.
- Ll is the value of Llp and Lls’ (which are assumed equal).

Having measured the short circuit input inductance to be 43nH, it is distributed equally over Llp and Lls’ so Ll is entered as 22nH.

Above is a screenshot of the Simsmith model. Block D1 is used for data entry to supply some values direct and calculated to the following blocks.

Tfmr is the model of the transformer as shown in the diagram earlier.

Above is a plot of the measured total leakage inductance over 1-30MHz.

Above is a plot of calculated 1-k where k is the flux coupling factor. Again the measured leakage inductance and winding inductances show that k is not independent of frequency, and 1-k (which determines leakage inductance in a coupled inductor model) varies over more than 2:1 range in this example. The graph demonstrates that models that are based on an assumption that k and 1-k are independent of frequency are flawed.

Above is the modelled VSWR response of the compensated transformer on a nominal load. It is very good from 3.5-30MHz.

Above, drilling down on more detail, the \(Loss=10 log \frac{PowerIn}{PowerOut}\) curve is very good. Maximum loss is at about 4MHz, and at 0.06dB loss @ 7.0MHz means that 98.6% of the transformer input power power reaches the transformer output terminals, the deficit being lost mostly in heating the ferrite core.

So, in contrast to the FT240-43 2t:14t transformer:

- ΣA/l is nearly nine times that of the FT240 core, so fewer turns are required for similar core loss;
- shorter winding length helps to reduce flux leakage;
- lower leakage flux improves VSWR bandwidth;
- smaller cores can dissipate less heat;
- reducing core loss reduces the need to dissipate as much heat; and
- compensation capacitor assumes Q of silver mica, the appropriate choice for a transmitting application;

The transformer in free air can probably dissipate around 2W continuous, an at 4MHz where transformer loss is 1.35%, continuous power rating would be 148W (200Ω load, free air). Of course an enclosure is likely to reduce power rating.

Note that leakage inductance is sensitive to the diameter of conductors and the spacing relative to other conductors, so changing the wire conductor diameter and insulation diameter, and wire to wire spacing all roll into changes in leakage inductance. For broadband performance, the goal is least leakage inductance.

Try changing model parameters in the sample model (link below), change mix type, measure the leakage inductance for some different winding configurations and use it.

If you have heard online experts advising the #43 mix is not suitable for this type of application, and that you should use something else… try something else in the model… if you can find a binocular of this size in a more suitable material.

The model input value aol is the core geometry ΣA/l (m) and can be calculated from dimensions using Calculate ferrite cored inductor – rectangular cross section. Some datasheets give ΣA/l or ΣA/l in various units which can be inverted / scaled as necessary. Calculate ferrite cored inductor (from Al) can calculate ΣA/l (m) from Al.

The model does not give a definitive design, but it does help to explore the effects of magnetising admittance and leakage inductance on VSWR bandwidth, loss etc.

Sample Simsmith model for download: EFHW-2843009902-43-2020-3-6k.7z . (Compressed with 7zip.)

]]>This article models the transformer on a nominal load, being \(Z_l=n^ 2 50 \;Ω\). Real EFHW antennas operated at their fundamental resonance and harmonics are not that simple, so keep in mind that this level of design is but a pre-cursor to building a prototype and measurement and tuning with a real antenna.

Above is the prototype transformer measured using a nanoVNA, the measurement is of the inductance at the primary terminals with the secondary short circuited.

The prototype transformer follows the very popular design of a 2:14 turns transformer with the 2t primary twisted over the lowest 2t of the secondary, and the winding distributed in the Reisert style cross over configuration.

The winding layout used in the prototype is that recommended at 10/(15)/20/40 Mini End fed antenna kit, 100 Watt 1:49 impedance transformer .

Above is a plot of the equivalent series primary inductance of the prototype transformer with short circuit secondary calculated from s11 measured with a nanoVNA from 1-31MHz. Note that the inductance is fairly independent of frequency, rising a little at the high frequency end probably due to effects of distributed capacitance and self resonance. This suggests that leakage flux is for the most part not immersed in the ferrite core, and it provides hints as to how to minimise it.

Note that since the inductance of the primary and secondary are frequency dependent (by virtue of the ferrite characteristic), and that leakage inductance is relatively independent of frequency (see above), that the flux coupling coefficient k is frequency dependent, and making it constant is not a very good model at these frequencies.

It might appear that k is fairly independent of freq, but 1-k is not, and it is 1-k that is used to evaluate leakage inductance in the k based approach, so it delivers a poor estimate of leakage inductance when the magnetising inductance is frequency dependent (as it is likely to be with ferrite).

It can be seen above that 1-k varies over a 2:1 range in this model, which would drive a 2:1 variation in leakage inductance… when leakage inductance is almost constant (see the earlier chart).

Above is the equivalent circuit used to model the transformer. The transformer is replaced with an ideal 1:n transformer, and all secondary side values are referred to the primary side.

- Secondary side leakage inductance Lls is divided by n^2 to obtain the value primary referred leakage inductance in the circuit diagram.
- Cse is an equivalent shunt capacitance to partially model self resonance effects.
- Bm is the magnetising susceptance (calculated from other parameters).
- Gm is the magnetising conductance (calculated from other parameters).
- Llp is the primary side leakage inductance.
- Ccomp is a compensation capacitance.

A Simsmith model was built to implement the transformer model above.

- Complex core permeability is captured from a permeability data file.
- np is the number of turns on the primary.
- ratio is the turns ratio.
- cores is the number of cores in a stack.
- cse is Cse per the circuit diagram.
- Ll is the value of Llp and Lls’ (which are assumed equal).

Having measured the short circuit input inductance to be 360nH, it is distributed equally over Llp and Lls’ so Ll is entered as 180nH.

Above is a screenshot of the Simsmith model. Block D1 is used for data entry to supply some values direct and calculated to the following blocks.

Tfmr is the model of the transformer as shown in the diagram earlier.

Above is the modelled VSWR response of the compensated transformer on a nominal load. It is not brilliant, but might be acceptable to many users.

Above, drilling down on more detail, the \(Loss=10 log \frac{PowerIn}{PowerOut}\) curve is troubling. 1dB loss @ 7.0MHz means that only 74% of the transformer input power power reaches the transformer output terminals, the deficit being lost mostly in heating the ferrite core. Of course you transmitter might not develop its rated power into the load that it sees, there could be a further reduction in power output.

So despite its popularity, this is an appalling design. It has high loss due to insufficient turns, and high leakage inductance due to winding layout and high turns. Acceptable designs are a compromise between bandwidth and loss for a give core, and small is beautiful from the transmission parameters, but not for power handling.

Try changing model parameters in the sample model (link below), change mix type, measure the leakage inductance for some different winding configurations and use it.

The model input value aol is the core geometry ΣA/l (m) and can be calculated from dimensions using Calculate ferrite cored inductor – rectangular cross section. Some datasheets give ΣA/l or ΣA/l in various units which can be inverted / scaled as necessary. Calculate ferrite cored inductor (from Al) can calculate ΣA/l (m) from Al.

The model does not give a definitive design, but it does help to explore the effects of magnetising admittance and leakage inductance on VSWR bandwidth, loss etc.

Sample Simsmith model for download: EFHW-5943003801-43-2020-2-14xk.7z . (Compressed with 7zip.)

]]>This article models the transformer on a nominal load, being \(Z_l=n^ 2 50 \;Ω\). Real EFHW antennas operated at their fundamental resonance and harmonics are not that simple, so keep in mind that this level of design is but a pre-cursor to building a prototype and measurement and tuning with a real antenna.

The prototype transformer follows the very popular design of a 2:16 turns transformer with the 2t primary twisted over the lowest 2t of the secondary, and the winding distributed in the Reisert style cross over configuration.

Above is a plot of the equivalent series impedance of the prototype transformer with short circuit secondary calculated from s11 measured with a nanoVNA from 1-31MHz. Note that it is almost entirely reactive, and the reactance is almost proportional to frequency suggesting close to a constant inductance.

Above is a plot of the equivalent series primary inductance of the prototype transformer with short circuit secondary calculated from s11 measured with a nanoVNA from 1-31MHz. Note that the inductance is fairly independent of frequency, rising a little at the high frequency end probably due to effects of distributed capacitance and self resonance. This suggests that leakage flux is for the most part not immersed in the ferrite core, and it provides hints as to how to minimise it.

Note that since the inductance of the primary and secondary are frequency dependent (by virtue of the ferrite characteristic), and that leakage inductance is relatively independent of frequency (see above), that the flux coupling coefficient k is frequency dependent, and making it constant is not a very good model at these frequencies.

Above is the prototype transformer measured using a LCR meter, the measurement 335nH @ 100kHz is of the inductance at the primary terminals with the secondary short circuited.

Above is the equivalent circuit used to model the transformer. The transformer is replaced with an ideal 1:n transformer, and all secondary side values are referred to the primary side, eg \(L_{ls}^\prime=\frac{L_{ls}}{n^2}\).

- Secondary side leakage inductance Lls is divided by n^2 to obtain the value primary referred leakage inductance in the circuit diagram.
- Cse is an equivalent shunt capacitance to partially model self resonance effects.
- Bm is the magnetising susceptance (calculated from other parameters).
- Gm is the magnetising conductance (calculated from other parameters).
- Llp is the primary side leakage inductance.
- Ccomp is a compensation capacitance.

A Simsmith model was built to implement the transformer model above.

- Complex core permeability is captured from a permeability data file.
- np is the number of turns on the primary.
- ratio is the turns ratio.
- cores is the number of cores in a stack.
- cse is Cse per the circuit diagram.
- Ll is the value of Llp and Lls’ (which are assumed equal).

Having measured the short circuit input inductance to be 350nH, it is distributed equally over Llp and Lls’ so Ll is entered as 175nH.

Above is a screenshot of the Simsmith model. Block D1 is used for data entry to supply some values direct and calculated to the following blocks.

Tfmr is the model of the transformer as shown in the diagram earlier.

Above is the modelled VSWR response of the compensated transformer on a nominal load. It is not brilliant, but might be acceptable to many users.

Above, drilling down on more detail, the \(Loss=10 log \frac{PowerIn}{PowerOut}\) curve is troubling. 1dB loss @ 3.5MHz means that only 74% of the transformer input power power reaches the transformer output terminals, the deficit being lost mostly in heating the ferrite core. Of course you transmitter might not develop its rated power into the load that it sees, there could be a further reduction in power output.

So despite its popularity, this is an appalling design. It has high loss due to insufficient turns, and high leakage inductance due to winding layout and high turns. Acceptable designs are a compromise between bandwidth and loss for a give core, and small is beautiful from the transmission parameters, but not for power handling.

Try changing model parameters in the sample model (link below), change mix type, measure the leakage inductance for some different winding configurations and use it.

The model input value aol is the core geometry ΣA/l (m) and can be calculated from dimensions using Calculate ferrite cored inductor – rectangular cross section. Some datasheets give ΣA/l or ΣA/l in various units which can be inverted / scaled as necessary. Calculate ferrite cored inductor (from Al) can calculate ΣA/l (m) from Al.

The model does not give a definitive design, but it does help to explore the effects of magnetising admittance and leakage inductance on VSWR bandwidth, loss etc.

Sample Simsmith model for download: EFHW-5943003801-43-2020-2-16x.7z . (Compressed with 7zip.)

]]>References without any qualification surely imply a recommendation.

In the same thread, Roger Need compared his measurement of a FT50-43 with Calculate ferrite cored inductor (from Al) (one of a set of related calculators), and Ferrite permeability interpolations.

Above, his calculation reconciles well with measurement at 3.6MHz.

Above is the same scenario calculated with VK3CPU’s calculator. An impressive GUI, if you like that sort of thing… and people do. Note the values at 3.6MHz in the cursor callout.

The value L is 7.06µH and it varies with frequency, but what does L mean. The notes state:

Tapping on a data point will display the parameters for a single frequency.

L(μH) : Inductance in microhenries.

Roger’s measured impedance @ 3.6MHz implies an equivalent series inductance (Ls in my calculators) of 3.965µH.

Though several posters to the thread have discussed VK3CPU’s calculator and the merits of it, it seems no one tried to reconcile it with Roger’s or any other measurements.

Wait a minute, v0.8 just released gives different values (including data for old #43 mix).

Now L=4.09µH which reconciles with Roger’s measurement.

]]>This article explains a little of the detail behind the graph.

The graph is based on a series of NEC-4.2 models of the loop in ground antenna. Key model parameters are:

- 3m a side;
- ‘average’ soil (σ=0.005, εr=13);
- depth=0.02m; and
- frequency 0.5 to 10MHz in 0.1MHz increments.

The models were scripted by a PERL script, and the output parsed with a Python script to extract feed point Z, structure efficiency, and average power gain (corrected to 4πsr).

The summarised NEC data was imported into a spreadsheet and an approximate model of the system built, comprising:

- Receiver input impedance 50+j0Ω;
- a length of transmission line (10m of Belden 8215 RG6/U);
- an ideal transformer (4:1);
- source impedance derived from the NEC data.

Calculation includes:

- transmission line loss and impedance transformation;
- transformer assumed ideal plus an allowance for transformer loss (1dB);
- mismatch loss; and
- average antenna gain.

Above is an extract of the spreadsheet.

Mismatch loss is an important element of the system behavior. A convenient place at which to calculate mismatch loss is the feed point of the loop in ground.

Above is a plot of the loop feed point impedance, the source impedance in the receive scenario.

Above is a plot of the loop load impedance, the receiver impedance transformed by transmission line and transformer. The varying impedance is a result of using 75Ω line.

The combination of these allows us to calculate mismatch loss.

Above is a plot of the calculated mismatch loss which must be added in to the system gain model.

From the system model, and an estimate of ambient noise from ITU-R P.372-14, we can calculate SND.

Above is a plot of SND.

Note that P.372-14 is based on a survey with short vertical monopole antennas, so it is likely to overestimate noise received by a horizontally polarised antenna (and therefore the SND estimate will be low).

Antenna performance is sensitive to soil parameters, especially those close to the surface and subject to variation with recent rainfall etc.

This is after all a feasibility study, and within acceptable uncertainty, the antenna system would seem to be feasible for low HF and even 160m receive.

]]>Let’s take ambient noise as Rural precinct in ITU-P.372-14.

An NEC-4.2 model of the 3m a side LiG gives average gain -37.18dBi. An allowance of 2.7dB of feed loss covers actual feed line loss and mismatch loss.

Above, calculated SND is 0.9dB. For this scenario (ambient noise and antenna system), the receiver S/N is 0.6dB worse than the off-air or intrinsic S/N ration. For Residential precinct ambient noise, SND is less at 0.3dB.

The above graph shows the system behavior over 0.5-10MHz, it is a combination of the effects of noise distribution; antenna gain; mismatch; transformer and feedline losses; and receiver internal noise.